Vertical adaptive antenna array for a discrete multitone spread spectrum communications system

ABSTRACT

Two or more antenna elements are arranged in the vertical direction to give vertical spatial adaptivity to a wireless discrete multitone spread spectrum communications system. The system is based on a combination of Discrete Multitone Spread Spectrum (DMT-SS) and multi-element adaptive antenna array technologies. This enables the automatic positioning of a beam in the vertical direction to position nulls where interferers are located on the same azimuth but are separated in elevation.

[0001] The invention disclosed herein is related to the copending U.S.patent application by S. Alamouti, D. Michaelson, E. Casas, E. Hoole, G.Veintimilla, H. Zhang, M. Hirano, and P. Poon, entitled “Method forFrequency Division Duplex Communications in a Personal Wireless AccessNetwork”, serial. No. ______, filed on the same day as the instantpatent application, assigned to AT&T, and incorporated herein byreference.

FIELD OF THE INVENTION

[0002] This invention involves improvements to a wireless a discretemultitone spread spectrum communications system.

BACKGROUND OF THE INVENTION

[0003] Wireless communications systems, such as cellular and personalcommunications systems, operate over limited spectral bandwidths. Theymust make highly efficient use of the scarce bandwidth resource toprovide good service to a large population of users. Code DivisionMultiple Access (CDMA) protocol has been used by wireless communicationssystems to efficiently make use of limited bandwidths. The protocol usesa unique code to distinguish each user's data signal from other users'data signals. Knowledge of the unique code with which any specificinformation is transmitted, permits the separation and reconstruction ofeach user's message at the receiving end of the communication channel.

[0004] The personal wireless access network (PWAN) system describedbelow, uses a form of the CDMA protocol known as discrete multitonespread spectrum ( DMT-SS ) to provide efficient communications between abase station and a plurality of remote units. (The term “discretemultitone stacked carrier (DMT-SS)” also refers to this protocol.) Inthis protocol, the user's data signal is modulated by a set of weighteddiscrete frequencies or tones. The weights are spreading codes thatdistribute the data signal over many discrete tones covering a broadrange of frequencies. The weights are complex numbers with the realcomponent acting to modulate the amplitude of a tone while the complexcomponent of the weight acts to modulate the phase of the same tone.Each tone in the weighted tone set bears the same data signal. Pluralusers at the transmitting station can use the same tone set to transmittheir data, but each of the users sharing the tone set has a differentset of spreading codes The weighted tone set for a particular user istransmitted to the receiving station where it is processed withdespreading codes related to the user's spreading codes, to recover theuser's data signal. For each of the spatially separated antenna arrayelements at the receiver, the received multitone signals are transformedfrom time domain signals to frequency domain signals. Despreadingweights are assigned to each frequency component of the signals receivedby each antenna array element. The values of the despreading weights arecombined with the received signals to obtain an optimized approximationof individual transmitted signals characterized by a particularmultitone set and transmitting location. The PWAN system has a total of2560 discrete tones (carriers) equally spaced in 8 MHZ of availablebandwidth in the range of 1850 to 1990 MHZ. The spacing between thetones is 3.125 kHz. The total set of tones are numbered consecutivelyfrom 0 to 2559 starting from the lowest frequency tone. The tones areused to carry traffic messages and overhead messages between the basestation and the plurality of remote units. The traffic tones are dividedinto 32 traffic partitions, with each traffic channel requiring at leastone traffic partition of 72 tones.

[0005] In addition, the PWAN system uses overhead tones to establishsynchronization and to pass control information between the base stationand the remote units. A Common Link Channel (CLC) is used by the base totransmit control information to the Remote Units. A Common AccessChannel (CAC) is used to transmit messages from the Remote Unit to theBase. There is one grouping of tones assigned to each channel. Theseoverhead channels are used in common by all of the remote units whenthey are exchanging control messages with the base station.

[0006] In the PWAN system, Time Division Duplexing (TDD) is used by thebase station and the remote unit to transmit data and controlinformation in both directions over the same multi-tone frequencychannel. Transmission from the base station to the remote unit is calledforward transmission and transmission from the remote unit to the basestation is called reverse transmission. The time between recurrenttransmissions from either the remote unit or the base station is the TDDperiod. In every TDD period, there are four consecutive transmissionbursts in each direction. Data is transmitted in each burst usingmultiple tones. The base station and each remote unit must synchronizeand conform to the TDD timing structure and both the base station andthe remote unit must synchronize to a framing structure. All remoteunits and base stations must be synchronized so that all remote unitstransmit at the same time and then all base stations transmit at thesame time. When a remote unit initially powers up, it acquiressynchronization from the base station so that it can exchange controland traffic messages within the prescribed TDD time format. The remoteunit must also acquire frequency and phase synchronization for theDMT-SS signals so that the remote is operating at the same frequency andphase as the base station.

[0007] The PWAN system provides for base station beam steering in thehorizontal plane. Interference sources that are located in substantiallythe same horizontal plane as the base station antenna array can havetheir effects reduced by steering the received sensitivity directionaway from that source. A corresponding horizontal shift can be made inthe transmitted beam direction to avoid creating interference at thelocation of the interfering source. In effect, nulls are steered ontothe interfering sources. However, a problem arises when there areinterfering sources located in the same radial direction from the basestation as a remote unit but are separated in the vertical plane. Suchsources cannot be adequately minimized by the PWAN system as the beamand null resolution attained in the horizontal plane is insufficient.

SUMMARY OF THE INVENTION

[0008] This problem is solved, in accordance with the invention, byproviding two or more antenna arrays, here called subarrays arranged inthe vertical direction to give spatial adaptivity in the vertical planeto the wireless discrete multitone spread spectrum communicationssystem. The PWAN system is based on a combination of Discrete MultitoneSpread Spectrum (DMT-SS) and multi-element adaptive antenna arraytechnologies. The inventors have discovered that the spatial andspectral processing of the DMT-SS signals received or transmitted fromthe base station antenna array is independent of the type of antennaarray utilized. This processing maximized the overallsignal-to-interference level of a base station's coverage area. Usingvertically-separated subarrays (composed of horizontal antenna elements)enables the base station to position beams on remote units in thevertical plane as well as position nulls on interferers located at thesame azimuth angle as the remote but which are separated in elevationangle. This additional vertical plane adaptivity will enhance the PWANsystem's overall performance and improve the signal-to-interferencelevel.

[0009] The invention is a highly bandwidth-efficient communicationsmethod to enable vertical and horizontal receive beam steering, thatincludes the following steps. The base station receives a first spreadsignal at a base station having a multi-subarray antenna array with afirst plurality of subarrays arranged in a spaced vertical direction anda second plurality of subarray elements arranged in a spaced horizontaldirection. The first spread signal comprises a first data signal spreadover a plurality of discrete tones in accordance with a remote spreadingcode assigned to a remote unit for a first time period. The base stationadaptively despreads the received signal by using first despreadingcodes that are based on the characteristics of the received signals atthe first plurality of subarrays of the array. The vertical displacementof the subarrays enables the base station to perform vertical, receivebeam steering. In addition, the base station adaptively despreads thesignal received by using second despreading codes that are based on thecharacteristics of the received signals at the second plurality ofsubarray elements of the array. The horizontal displacement of thesubarray elements enables the base station to perform horizontal,receive beam steering.

[0010] The invention also enables transmit beam steering in the verticaland horizontal directions. The method includes the following additionalsteps The base station spreads a second data signal with first spreadingcodes derived from the first despreading codes, that distributes thesecond data signal over a plurality of discrete tones and the firstplurality subarrays of the array, forming a first spectrally spreadsignal that is spatially spread vertically. The vertical displacement ofthe subarray enables the base station to perform vertical, transmit beamsteering. The base station also spreads the second data signal withsecond spreading codes derived from the second despreading codes, thatdistributes the second data signal over the plurality of discrete tonesand the second plurality subarray elements of the array, forming asecond spectrally spread signal that is spatially spread horizontally.The horizontal displacement of the subarray enables the base station toperform horizontal, transmit beam steering. The base station thentransmits the first and second spread signals during a second timeperiod.

[0011] The invention has advantageous applications in the field ofwireless communications, such as cellular communications or personalcommunications, where bandwidth is scarce compared to the number of theusers and their needs. Such applications may be effected in mobile,fixed, or minimally mobile systems.

BRIEF DESCRIPTION OF THE DRAWINGS

[0012] In the drawings:

[0013]FIG. 1A shows a linear antenna array at a base station where thesignal processor distinguishes two discrete monotone signals receivedfrom two remote units that are placed far from one another.

[0014]FIG. 1B shows a vertical and horizontal, two dimensional antennaarray at the base station, in accordance with the invention, where thesignal processor distinguishes two discrete monotone signals receivedfrom the two remote units X and Y that are separated from one another inthe vertical plane

[0015]FIG. 1C shows the linear antenna array at the base station of FIG.1A, where the signal processor processes two discrete monotone signalsfor transmission to two remote units that are displaced from oneanother.

[0016]FIG. 1D shows the vertical and horizontal, two dimensional antennaarray at the base station of FIG. 1B, in accordance with the invention,where the signal processor processes two discrete monotone signals fortransmission to two remote units that are separated from one another inthe vertical plane

[0017]FIG. 2 shows an alternate embodiment of the vertical andhorizontal, two dimensional antenna array at the base station of FIG. 1Band FIG. 1D, in accordance with the invention, the elements of eachsubarray are disposed around a cylinder.

[0018]FIG. 3A is a tutorial diagram illustrating an example of purespectral diversity, showing how a receiver distinguishes two sets ofdiscrete multitone signals from two transmitters that are placed closeto one another, in accordance with the PWAN system.

[0019]FIG. 3B is a tutorial diagram illustrating an example of purespatial diversity, showing how a receiver distinguishes two discretemonotone signals from two transmitters that are placed far from oneanother, in accordance with the PWAN system.

[0020]FIG. 3C is a tutorial diagram illustrating an example of bothspectral and spatial diversity, showing how a receiver distinguishes twodiscrete multitone signals from two transmitters that are placed farfrom one another, in accordance with the PWAN system.

[0021]FIG. 3D is a high-level schematic representation of animplementation of the PWAN system in a fixed wireless communicationsystem.

[0022]FIG. 3E is a simplified representation of multitone transmission.

[0023]FIG. 3F is a simplified representation of the use of a discretemultitone spread spectrum signal format.

[0024]FIG. 4 is a simplified representation of the matrix formalism usedin an implementation of the PWAN system.

[0025]FIG. 5 is a simplified representation of the matrix formalism,used in an implementation of the PWAN system, that includes the effectsof channel response.

[0026]FIG. 6 is a simplified representation of DMT-SS using an exemplaryhigher order QAM modulation format.

[0027]FIG. 7 is a timing diagram that illustrates the general timedivision duplex signal and protocol used in an embodiment of the PWANsystem.

[0028]FIG. 8 is a signal processing flow diagram that depicts the mainsignal processing steps used in an embodiment of the PWAN system toprovide for high bandwidth efficiency.

[0029]FIG. 9 is a signal processing flow diagram that illustrates amethod used to spread the encoded carrier signal.

[0030]FIG. 10 is a three-dimensional plot of the signal to interferenceplus noise ratio versus code weights and spatial weights applied to thetransmitted and received signals.

[0031]FIG. 11 is a perspective cut away view showing an embodiment of abase station antenna.

[0032]FIG. 12 is a perspective cut away view showing a second embodimentof a base station antenna.

[0033]FIG. 13 graphically depicts the null steering aspect of thepresent PWAN system.

[0034]FIG. 14 is a schematic representation of an inverse frequencychannelized spreader implementation.

[0035]FIG. 15 is a schematic representation of a frequency channelizeddespreader implementation.

[0036]FIG. 16 is a plot of antenna gain versus angular direction.

[0037]FIG. 17 is a highly simplified block diagram that illustrates oneparticular application of the highly bandwidth-efficient communicationsnetwork of the present PWAN system.

[0038]FIG. 18 is a list of the possible operational frequency bands of aspecific embodiment of the PWAN system.

[0039]FIG. 19 shows the RF Band/Sub-band organization of the airlink ofa specific embodiment of the PWAN system.

[0040]FIG. 20 shows the tones within each sub-band of a specificembodiment of the PWAN system

[0041]FIG. 21 shows the traffic partitions in a specific embodiment ofthe PWAN system

[0042]FIG. 22 shows the tone mapping to the ith traffic partition

[0043]FIG. 23 shows the overhead tone Mapping to Channels for the ithSub-band Pair

[0044]FIG. 24 shows the Division of Tone Space to Traffic and OverheadTones

[0045]FIG. 25 shows the time Division Duplex format for Base and RemoteUnit Transmissions

[0046]FIG. 26 shows Details of the Forward and Reverse Channel TimeParameters

[0047]FIG. 27 shows the TDD Parameter Values

[0048]FIG. 28 shows the Physical Layer Framing Structure

[0049]FIG. 29 shows the Phase A Sub-band Pair Assignment Within a Cell

[0050]FIG. 30 shows the Phase-A Sub-band Pair Assignment Across Cells

[0051]FIG. 31 is a Functional Block Diagram for the Upper Physical Layerof Base Transmitter for High Capacity Mode

[0052]FIG. 32 is a Data Transformation Diagram for the High CapacityForward Channel Transmissions

[0053]FIG. 33 is a Functional Block Diagram for the Upper Physical Layerof Base Transmitter for Medium Capacity Mode

[0054]FIG. 34 is a Data Transformation Diagram for the Medium CapacityForward Channel Transmissions

[0055]FIG. 35 is a Functional Block Diagram for the Upper Physical Layerof Base Transmitter for Low Capacity Mode

[0056]FIG. 36 is a Data Transformation Diagram for the Low CapacityForward Channel Transmissions

[0057]FIG. 37 is a representation of the Triple DES Encryption Algorithm

[0058]FIG. 38 depicts a Feed Forward Shift Register Implementation ofRate 3/4, 16PSK Trellis Encoder for High Capacity Mode

[0059]FIG. 39 depicts a Feed Forward Shift Register Implementation ofRate 3/4, 16QAM Trellis Encoder for High Capacity Mode

[0060]FIG. 40 shows the Signal Mappings for Rate 3/4, 16QAM and 16PSKTrellis Encoding Schemes Employed in High Capacity Mode

[0061]FIG. 41 shows the Signal Mappings for Rate 3/4, Pragmatic 16 QAMand 16 PSK Trellis Encoding Schemes Employed in High Capacity Mode

[0062]FIG. 42 depicts a Feed Forward Shift Register Implementation ofRate 2/3, 8PSK Trellis Encoder for Medium Capacity Mode

[0063]FIG. 43 depicts a Feed Forward Shift Register Implementation ofRate 2/3 8QAM Trellis Encoder for Medium Capacity Mode

[0064]FIG. 44 shows the Signal Mappings for Rate 2/3, 8QAM and 8PSKTrellis Encoding Schemes Employed in Medium Capacity Mode

[0065]FIG. 45 shows the Signal Mappings for Rate 2/3, 8QAM and 8PSKTrellis Encoding Schemes Employed in Medium Capacity Mode

[0066]FIG. 46 depicts a Feed Forward Shift Register Implementation ofRate << Convolutional Encoder for Low Capacity Mode

[0067]FIG. 47 shows the Signal Mapping for Rate <<, QPSK PragmaticTrellis Encoding Scheme Employed in Low Capacity Mode

[0068]FIG. 48 shows the Gray-Coded Mapping for Rate <<, QPSK PragmaticTrellis Encoding Scheme Employed in Low Capacity Mode

[0069]FIG. 49 shows the Base Mapping of Elements of Received WeightVectors to Antenna Elements and Tones

[0070]FIG. 50 is a Block Diagram Representation of CLC Physical LayerFormat

[0071]FIG. 51 shows the QPSK Signal Mapping for the CLC Channel

[0072]FIG. 52 is a representation of the CLC Interleaving Rule

[0073]FIG. 53 shows the Tone Mapping of (4×4) Interleaved MatrixElements

[0074]FIG. 54 is a Block Diagram Representation of BRC Physical LayerFormat

[0075]FIG. 55 shows the Tone Mapping of the (4×4) Interleaved MatrixElements

[0076]FIG. 56 is a representation of a Broadcast Channel Beam Sweep

[0077]FIG. 57 is a Functional Block Diagram of the Upper Physical Layerof Remote Unit Transmitter for High Capacity Mode

[0078]FIG. 58 is a Data Transformation Diagram for the High CapacityReverse Channel Transmissions

[0079]FIG. 59 is a Functional Block Diagram for the Upper Physical Layerof Remote Unit Transmitter for Medium Capacity Mode

[0080]FIG. 60 is a Data Transformation Diagram for the Medium CapacityReverse Channel Transmissions

[0081]FIG. 61 is a Functional Block Diagram for the Upper Physical Layerof Remote Unit Transmitter for Low Capacity Mode

[0082]FIG. 62 is a Data Transformation Diagram for the Low CapacityReverse Channel Transmissions

[0083]FIG. 63 shows the Remote Unit Tone Mapping of Received WeightVector Elements

[0084]FIG. 64 is a Block Diagram Representation of the CAC PhysicalLayer Format

[0085]FIG. 65 shows the BPSK Signal Mapping for the CAC Channel

[0086]FIG. 66 depicts the CAC Interleaving Rule

[0087]FIG. 67 shows the Tone Mapping of the (8×2) Interleaved MatrixElements

[0088]FIG. 68 is a Functional Block Diagram for the Lower Physical Layerof Base Transmitter

[0089]FIG. 69 shows Tone Mapping into DFT Bins

[0090]FIG. 70 shows Tone Mapping into DFT Bins

[0091]FIG. 71 is a block diagram that illustrates the main structuraland functional elements of the bandwidth on demand communicationsnetwork of the present PWAN system.

[0092]FIG. 72 is a functional block diagram that illustrates the mainfunctional elements of the high bandwidth remote access station.

[0093]FIG. 73 is a functional block diagram that shows the mainfunctional components of the high bandwidth base station.

[0094]FIG. 74 is an overall system schematic block diagram that showsthe main structural and functional elements of one implementation of thehighly bandwidth-efficient communication system in greater detail.

[0095]FIGS. 75A depict the digital architecture within an exemplaryremote access terminal.

[0096]FIGS. 75B depict the digital architecture within an exemplaryremote access terminal.

[0097]FIG. 76 is a software block diagram that indicates the generalprocessing steps performed by each of the digital signal processingchips within the digital signal processing architecture of FIGS. 75A and75B.

[0098] FIGS. 77A-77D are block diagrams that show in detail the digitalarchitecture of the LPA cards of FIGS. 75A and 75B.

[0099] FIGS. 78A-78C are block diagrams that detail the digitalarchitecture used to support the main digital signal processing chips onthe interface card of FIGS. 75A and 75B.

[0100] FIGS. 79A-79D are a schematic block diagram that depicts theoverall digital signal processing architectural layout within anexemplary base station of the present PWAN system.

[0101]FIG. 80 is a schematic block diagram showing a dual band radiofrequency transceiver that may advantageously be used in the highbandwidth remote access station shown in FIG. 74.

[0102]FIG. 80A is a schematic block diagram showing the main internalfunctional elements of the synchronization circuitry shown in FIG. 80.

[0103]FIG. 81 is a schematic block diagram depicting a dual band radiofrequency transceiver that may advantageously be implemented within thehigh bandwidth base station shown in FIG. 74.

[0104]FIG. 81A is a simplified schematic block diagram showing the maininternal components of the frequency reference circuit shown in FIG. 81.

[0105]FIG. 82 is a schematic block diagram of a dual band radiofrequency transmitter of a type that may advantageously be implementedwithin a base station constructed in accordance with the present PWANsystem.

[0106]FIG. 83 depicts the bandwidth allocation method performed by thebandwidth demand controller of FIG. 74.

[0107]FIG. 84A and FIG. 84B show an alternate embodiment of the PWANsystem, where the spectral processing and the spatial processing areseparated. The spatial weights are computed independently for eachcarrier frequency.

[0108]FIG. 85 is an illustrative flow chart of an embodiment of theadaptive solution of spectral and spatial weights.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

[0109] The invention is a highly bandwidth-efficient communicationsmethod to enable vertical and horizontal receive beam steering, thatincludes the following steps. The base station receives a first spreadsignal at a base station having a multi-subarray antenna array with afirst plurality of antenna elements arranged in a spaced verticaldirection and a second plurality of subarray elements arranged in aspaced horizontal direction. The first spread signal comprises a firstdata signal spread over a plurality of discrete tones in accordance witha remote spreading code assigned to a remote unit for a first timeperiod. The base station adaptively despreads the received signal byusing first despreading codes that are based on the characteristics ofthe received signals at the first plurality of subarrays of the array.The vertical displacement of the subarrays enables the base station toperform vertical, receive beam steering. In addition, the base stationadaptively despreads the signal received by using second despreadingcodes that are based on the characteristics of the received signals atthe second plurality of subarray elements of the array. The horizontaldisplacement of the subarray elements enables the base station toperform horizontal, receive beam steering.

[0110] The invention also enables transmit beam steering in the verticaland horizontal directions. The method includes the following additionalsteps. The base station spreads a second data signal with firstspreading codes derived from the first despreading codes, thatdistributes the second data signal over a plurality of discrete tonesand the first plurality of subarrays of the array, forming a firstspectrally spread signal that is spat;ally spread vertically. Thevertical displacement of the subarrays enables the base station toperform vertical, transmit beam steering. The base station also spreadsthe second data signal with second spreading codes derived from thesecond despreading codes, that distributes the second data signal overthe plurality of discrete tones and the second plurality of subarrayelements of the array, forming a second spectrally spread signal that isspatially spread horizontally. The horizontal displacement of thesubarray elements enables the base station to perform horizontal,transmit beam steering. The base station then transmits the first andsecond spread signals during a second time period.

[0111] The invention uses pilot tones to make successively more preciseestimates of the despreading codes. This process begins in aninitialization period, when the base station receives a pilot spreadsignal comprising a known data signal spread over a plurality ofdiscrete tones. The signal processor in the base station correlates theknown data signal from the pilot spread signal with a reference knowndata signal and forms the despreading codes. For the verticallydisplaced subarrays, a first despreading code is estimated that is basedon the characteristics of the received signals at the subarrays, where agiven element of the first despreading code corresponds to a given oneof the vertical antenna elements and a given one of the discrete tones.In a similar manner, for the horizontally displaced elements of eachsubarray, the signal processor correlates the known data signal from thepilot spread signal with the reference known data signal and forms thesecond despreading code that is based on the characteristics of thereceived signals at the subarray elements. A given element of thedespreading code corresponds to a given one of the subarray elements anda given one of the discrete tones.

[0112]FIG. 1A shows a linear antenna array at a base station where thesignal processor distinguishes two discrete monotone signals receivedfrom two remote units that are placed far from one another. Transmittingremote unit X receives an input data signal shown as a white data signalfrom a first sender. The first sender is assigned a unique code of “1”that is used by the encoder at remote unit X to encode the white datasignal. In this example, there is only one tone at remote unit X. Theencoder encodes the white data signal onto a single discrete frequencyor tone: Tone 1. The white data signal is copied onto the tone. Themagnitude of a code digit value is converted by the encoder into acorresponding magnitude phase delay in positioning the copy of the whitesignal onto the discrete tone. A phase delay corresponding to a codedigit value of 1 is shown as positioning the copy of the white signal onits discrete tone at the time T1. (This example introduces the reader tothe invention's feature of spectral spreading by portraying the phaseencoding of the signal as a time delay modulation.) FIG. 1A also showstransmitting remote unit Y receiving an input data signal shown as ablack data signal from a second sender. Remote unit Y is geographicallydistant from remote unit X. In this example, the second sender isassigned the same unique code of “1” at that for the first sender. Thus,the black data signal is also copied onto the first discrete tone 1. Aphase delay corresponding to a code digit value of 1 is shown aspositioning the copy of the black signal onto its discrete tone at thetime T1. Both remote units X and Y have used the same tone and phase totransmit their respective data. In this example, we rely on the diversegeographic locations of remote units X and Y to distinguish the signalstransmitted from the two remote units. This is an example of spatialdiversity. The two signals from remote unit X and remote unit Y arediverse because the phases of their tones will appear arrive atdifferent times at a receiver.

[0113] This spatial diversity is detected by the receiving base stationZ of FIG. 1A, in accordance with one aspect of the PWAN system. Thedirection of transmission from the remote units to the base station isreferred to herein as the reverse channel and occurs in the reverseinterval of a time division duplex (TDD) period. FIG. 1A shows thereceiving base station Z receiving the discrete monotone signals on itsfour antennas A, B, C, and D over reverse channels from the remote unitsX and Y in. The signals are processed by a signal processor computerusing the PWAN spatial and spectral despreading codes and stored in amemory. The memory at the receiving base station Z is organized so thateach bin is associated with one of the four antennas at the receivingbase station and with one tone out of a possible plurality of fourtones. If there were four possible tones, then each antenna A, B, C, andD would have four bins in the memory. As before, each bin is furtherdivided into four sub-bins for each of the four possible phases, T1, T2,T3, and T4. FIG. 1A shows how the bins and sub-bins in the memory ofbase station Z store the patterns of the white data received on the fourantennas A, B, C, and D from remote unit X. The base station computesupdated values for its despreading weights based on the most recent datareceived on the reverse channel, as is described below for the PWANsystem.

[0114]FIG. 1A also shows how the bins and sub-bins in the memory of basestation Z store the patterns of the black data received on the fourarray elements A, B, C, and D from remote unit Y. The signal processorat base station Z uses the process of spatial despreading, in accordancewith one aspect of the PWAN system, to distinguish the white data fromthe black data. The first user's unique code “1” and the relative phasedelays in the arrival of the white data to the four array elements A, B,C, and D is used here to form a first spatial despreading code toextract the white data form all of the signals received by arrayelements A, B, C, and D. The second user's unique code “1” and therelative phase delays in the arrival of the black data to the fourantennas A, B, C, and D is used here to form a second spatialdespreading code to extract the black data from all of the signalsreceived by element A.

[0115]FIG. 1B shows a vertical and horizontal, two dimensional antennaarray at the base station, in accordance with the invention, where thesignal processor distinguishes two discrete monotone signals receivedfrom the two remote units X and Y that are separated from one another inthe vertical plane. The remote unit X is at a relatively high elevationangle and remote unit Y is at a relatively low elevation angle. Bothremote units are positioned off-axis from the normal to the plane of theantenna array. In accordance with the invention, array elements A and Dhave been arranged to be vertically displaced and antenna elements B andC have been arranged to be horizontally displaced. Transmitting remoteunit X receives an input data signal shown as a white data signal from afirst sender. The first sender is assigned a unique code of “1” that isused by the encoder at remote unit X to encode the white data signal. Inthis example, there is only one tone at remote unit X. The encoderencodes the white data signal onto a single discrete frequency or tone:Tone 1. The white data signal is copied onto the tone. The magnitude ofa code digit value is converted by the encoder into a correspondingmagnitude phase delay in positioning the copy of the white signal ontothe discrete tone. A phase delay corresponding to a code digit value of1 is shown as positioning the copy of the white signal on its discretetone at the time T1. FIG. 1B also shows transmitting remote unit Yreceiving an input data signal shown as a black data signal from asecond sender. Remote unit Y is separated from remote unit X in thevertical plane. In this example, the second sender is assigned the sameunique code of “1” at that for the first sender. Thus, the black datasignal is also copied onto the first discrete tone 1. A phase delaycorresponding to a code digit value of 1 is shown as positioning thecopy of the black signal onto its discrete tone at the time T1. Bothremote units X and Y have used the same tone and phase to transmit theirrespective data. In this example, we rely on the diverse elevationangles of remote units X and Y to distinguish the signals transmittedfrom the two remote units. This is an example of vertical plane spatialdiversity. The two signals from remote unit X and remote unit Y arediverse because the phases of their tones will appear arrive atdifferent times at array elements A and D. Since the pair of horizontalantenna elements B and C are positioned midway in the vertical directionbetween vertical array elements A and D, the two signals from remoteunit X and remote unit Y will appear arrive at the pair B and C at adifferent time than when they arrive at vertically displaced arrayelements A and D. Since array element C is closer to the remote unitsthan is array element B, the signals from a given remote unit willarrive at C sooner than they will arrive at B. The delay in the arrivaltimes at the respective horizontal elements B and C gives the horizontalazimuth of the remote unit relative to the vertical plane passingperpendicularly through the axis of the horizontally displaced pair Band C. This two dimensional spatial diversity is detected by thereceiving base station Z of FIG. 1B in the same manner as it is for thelinear antenna array of FIG. 1A, in accordance with the invention.

[0116]FIG. 1B shows the receiving base station Z receiving the discretemonotone signals on its four array elements A, B, C, and D from theremote units X and Y. The signals are processed by the signal processorcomputer using the PWAN spatial and spectral despreading codes andstored in the memory in the same way as was described for the linearantenna array of FIG. 1A. The base station computes updated values forits despreading weights based on correlation estimates with the mostrecent data received on the reverse channel, as is described below forthe PWAN system. The memory at the receiving base station Z is organizedso that each bin is associated with one of the four array elements atthe receiving base station and with one tone out of a possible pluralityof four tones. As before, each bin is further divided into four sub-binsfor each of the four possible phases, T1, T2, T3, and T4. FIG. 1B showshow the bins and sub-bins in the memory of base station Z store thepatterns of the white data received on the four vertically andhorizontally separated array elements A, B, C, and D from remote unit X.FIG. 1B also shows how the bins and sub-bins in the memory of basestation Z store the patterns of the black data received on the fourspatially separated array elements A, B, C, and D from remote unit Y.The signal processor at base station Z uses the process of spatialdespreading, in accordance with one aspect of the PWAN system, todistinguish the white data from the black data. The first user's uniquecode “1” and the relative phase delays in the arrival of the white datato the four array elements A, B, C, and D is used here to form a firstspatial despreading code to extract the white data from all of thesignals received by array elements A, B, C, and D. The second user'sunique code “1” and the relative phase delays in the arrival of theblack data to the four array elements A, B, C, and D is used here toform a second spatial despreading code to extract the black data fromall of the signals received by array element A.

[0117] It can be seen by a comparison of FIG. 1A and FIG. 1B that thereis a similarity in the processing of spatially diverse signals of alinear antenna array and the processing of spatially diverse signalsfrom a vertical and horizontal two dimensional antenna array, when theyare encoded and decoded as has been described in accordance with oneaspect of the PWAN system.

[0118]FIG. 1C shows the linear antenna array at a base station of FIG.1A, where the signal processor processes two discrete monotone signalsfor transmission to two remote units that are displaced from oneanother. Transmission from the base station to the remote unit is calledthe forward channel and occurs in the forward interval of a timedivision duplex (TDD) period. The base station Z performs transmit beamforming, adjusting the phases of transmission from the respective arrayelements, to direct the signal to the desired remote unit. For example,the signal processor, using the PWAN spatial and spectral spreadingcodes, arranges the white data signals to be sent to remote unit X, bybeginning transmission from that array element that is the farthest fromremote unit X. The transmit spreading weights are a scaled version ofthe received despreading weights computed using the respective arrayelement inputs with each of the respective receive frequencies. Thespreading weights are computed and are applied to forward channel burstsafter a short delay. In this example, the signal processor arranges basestation's memory shown in FIG. 1C to have array element D transmittingwhite data at time T1, array element C transmitting at time T2, arrayelement B transmitting at time T3, and array element A transmitting attime T4. These relative phases for the respective transmitted white datasignals add together and form a beam that is directed to the remote unitX. A similar arrangement is produced by the signal processor, using thePWAN spatial and spectral spreading codes, to transmit the black data toremote station Y. In this example, the signal processor arranges basestation's memory shown in FIG. 1C to have array element A transmittingblack data at time T1, array element B transmitting at time T2, arrayelement C transmitting at time T3, and array element D transmitting attime T4. These relative phases for the respective transmitted black datasignals add together and form a beam that is directed to the remote unitY.

[0119]FIG. 1D shows the vertical and horizontal, two dimensional antennaarray at the base station of FIG. 1B, in accordance with the invention,where the signal processor processes two discrete monotone signals fortransmission to two remote units that are separated from one another inthe vertical plane. The base station Z performs transmit beam forming,adjusting the phases of transmission from the respective array elements,to direct the signal to the desired remote unit. The remote unit X is ata relatively high elevation angle and remote unit Y is at a relativelylow elevation angle. Both remote units are positioned off-axis from thenormal to the plane of the antenna array. In accordance with theinvention, array elements A and D have been arranged to be verticallydisplaced and array elements B and C have been arranged to behorizontally displaced. Since the pair of horizontally displaced arrayelements B and C are positioned midway in the vertical direction betweenvertically displaced array elements A and D, the two signals transmittedto remote unit X and remote unit Y will be timed for transmission fromthe pair B and C at a different instant than the time for transmissionfrom array elements A and D. Since array element C is closer to theremote units than is array element B, the signals transmitted to a givenremote unit are transmitted from element B at an earlier instant thanthey are transmitted from element C. The difference in the transmissioninstant from B and C corresponds to the horizontal azimuth of the remoteunit relative to the vertical plane passing perpendicularly through theaxis of the pair B and C. This two dimensional spatial diversity iscontrolled by PWAN spatial and spectral spreading codes at thetransmitting base station Z of FIG. 1D in the same manner as it is forthe linear antenna array of FIG. 1C, in accordance with the invention.

[0120] For example, the signal processor, using the PWAN spatial andspectral spreading codes, arranges the white data signals to be sent toremote unit X, by beginning transmission from that array element that isthe farthest from remote unit X. The transmit spreading weights are ascaled version of the received weights using the respective array inputswith each of the respective receive frequencies. The spreading weightsare applied to forward channel bursts after a short delay. In thisexample, the signal processor arranges base station's memory shown inFIG. 1D to have antenna D transmitting white data at time T1, element Btransmitting at time T2, element C transmitting at time T3, and elementD transmitting at time T4. These relative phases for the respectivetransmitted white data signals add together and form a beam that isdirected to the remote unit X. A similar arrangement is produced by thesignal processor, using the PWAN spatial and spectral spreading codes,to transmit the black data to remote station Y. In this example, thesignal processor arranges base station's memory shown in FIG. 1D to haveelement A transmitting black data at time T1, element B transmitting attime T2, element C transmitting at time T3, and element D transmittingat time T4. These relative phases for the respective transmitted blackdata signals add together and form a beam that is directed to the remoteunit Y.

[0121] It can be seen by a comparison of FIG. 1C and FIG. 1D that thereis a similarity in the processing of spatially diverse signals of alinear antenna array and the processing of spatially diverse signalsfrom a vertical and horizontal two dimensional antenna array, when theyare encoded and decoded as has been described in accordance with thePWAN system described below.

[0122]FIG. 2 shows an alternate embodiment of the vertical andhorizontal, two dimensional antenna array at the base station of FIG. 1Band FIG. 1D, in accordance with the invention, where the antennasubarrays are cylindrical. Subarrays elements A and D are arranged alongthe vertical axis and can be made up of subarrays arranged in acylindrical symmetry about the vertical Hs. Subarrays elements B and Care arranged in a cylindrical symmetry about the vertical axis. Thearrangement of elements A, B, C, and D provides the vertical andhorizontal, two dimensional antenna array in a similar manner as hasbeen described for the antenna elements of FIG. 1B and FIG. 1D. Thevertical arrangement of the subarrays elements A and D enables the basestation to perform vertical, receive and transmit beam steering. Thehorizontal orientation of the subarray elements B and C enables the basestation to perform horizontal, receive and transmit beam steering.Although FIG. 2 shows three cylindrical subarrays, the invention canalso be embodied as two cylindrical arrays, with the upper subarraycontaining both B and C type elements and the bottom subarray containboth C and B type elements. Although FIG. 2 shows three cylindricalarrays of subarrays, the invention can also be embodied as four or morecylindrical subarrays, with at least one of the vertically displacedsubarrays containing B and C type elements. The vertical spacing of thesubarrays A and D and the horizontal spacing of the subarray elements Band C can be chosen in accordance with the desired width of thetransmitted beam and the desired range of electromagnetic wavelengths tobe communicated

[0123] The resultant invention forms beams in both the horizontal andvertical planes. It solves the problem of forming a beam to a firstsubscriber when there is a second subscriber located on the samehorizontal azimuth but at a different elevation. The directed beam isformed using the PWAN system's spatial and spectral spreading anddespreading codes so as to maximize the signal to interference and noiseratio.

[0124] Detailed Description of the PWAN System

[0125] In what follows, aspects of the principles of the PWAN systemwill be discussed in a tutorial illustrating an example of pure spectraldiversity, an example of pure spatial diversity, and an example of mixedspectral and spatial diversity. This will be followed by a discussion ofthe PWAN system in a high-level overview that will include anexplanation of the waveform used in the practice of an aspect of thisPWAN system This will be followed by a description of more specific“details of the PWAN system,” and then by a detailed description of a“specific embodiment of the PWAN system.”

[0126] High Level Overview of the PWAN System

[0127] Introduction

[0128] This PWAN system is based, in part, on the realization that thereis an analogy between the mathematical description of beams formed bymulti-element adaptive, or phased, antenna arrays and the mathematicaldescription of signals that are formatted according to certain multipleaccess schemes, such as the exemplary DMT-SS Based on this realization,applicants have been able to simplify the calculations necessary when aplurality of multiple access techniques are combined. Using this PWANsystem, one may more effectively use a limited bandwidth region of theelectromagnetic spectrum to service a large number of users. Techniquesthat may be combined in accordance with the teachings of this PWANsystem include SDMA using multi-element antenna arrays, DMT-SS, andhigher order modulation formats such as higher order QAM.

[0129]FIG. 3A is a tutorial diagram illustrating an example of purespectral diversity, showing how a receiver distinguishes two sets ofdiscrete multitone signals from two transmitters that are placed closeto one another, in accordance with one aspect of the PWAN system.Transmitting station X receives an input data signal shown as a whitedata signal from a first sender. The first sender is assigned a uniquecode of “1234” that is used by the encoder at station X to encode thewhite data signal. In accordance with one aspect of the PWAN system, theencoder uses a discrete multitone spread spectrum protocol to encode thewhite data signal onto four discrete frequencies or tones: Tone 1, Tone2, Tone 3, and Tone 4. The white data signal is copied onto each of thefour tones. The first digit of the code is assigned to the firstdiscrete tone 1. The second digit of the code is assigned to the seconddiscrete tone 2, and so forth. The magnitude of a code digit value isconverted by the encoder into a corresponding magnitude phase delay inpositioning the copy of the white signal onto the discrete tone. A phasedelay corresponding to a code digit value of 1 is shown as positioningthe copy of the white signal on its discrete tone at the time T1. Aphase delay corresponding to a code digit value of 2 is shown aspositioning the copy of the white signal on its discrete tone one unitlater in time at T2, and so forth. The encoder at station X is shown inFIG. 3A as converting the first sender's unique code of “1234” intopositioning four copies of the white data at the phases T1, T2, T3, andT4 for Tone 1, Tone 2, Tone 3, and Tone 4, respectively. This process isspectral spreading and the unique code is a spectral spreading code.

[0130]FIG. 3A also shows transmitting station Y receiving an input datasignal shown as a black data signal from a second sender. The secondsender is assigned a unique code of “4321” that is used by the encoderat station Y to encode the black data signal. In accordance with oneaspect of the PWAN system, the encoder at station Y uses a discretemultitone spread spectrum protocol to encode the black data signal ontothe same four discrete frequencies or tones: Tone 1, Tone 2, Tone 3, andTone 4. The black data signal is copied onto each of the four tones. Thefirst digit of the code is assigned to the first discrete tone 1. Thesecond digit of the code is assigned to the second discrete tone 2, andso forth. The magnitude of a code digit value is converted by theencoder at station Y into a corresponding magnitude phase delay inpositioning the copy of the black signal onto the discrete tone. A phasedelay corresponding to a code digit value of 1 is shown as positioningthe copy of the black signal onto its discrete tone at the time T1. Aphase delay corresponding to a code digit value of 2 is shown aspositioning the copy of the black signal on its discrete tone one unitlater in time at T2, and so forth. The encoder at station Y is shown inFIG. 3A as converting the second sender's unique code of “4321” intopositioning four copies of the black data at the phases T1, T2, T3, andT4 for Tone 4, Tone 3, Tone 2, and Tone 1, respectively. This process isanother example of spectral spreading and the second user's unique codeis also a spectral spreading code.

[0131]FIG. 3A shows the transmitters at stations X and Y beingpositioned close to one another, so that the transmitted signals fromthem are not significantly different in their spatial characteristics.The transmitted signals from the two stations X and Y also have the samefour discrete frequencies or tones: Tone 1, Tone 2, Tone 3, and Tone 4.However, the transmitted signals from the two stations X and Y aredistinguishable by the phases of their tones, each having a patternrepresenting the unique code assigned to either the first or the secondsenders. This is an example of spectral diversity. The two signals fromstation X and station Y are diverse because the encoded phases of theirtones are diverse. This spectral diversity is detected by the receivingstation Z of FIG. 3A, in accordance with one aspect of the PWAN system.

[0132]FIG. 3A shows the receiving station Z receiving the discretemultitone signals on its antenna A from the stations X and Y. Thesignals are processed by a signal processor computer and stored in amemory. The memory at the receiving station Z is organized into sectionscalled bins. Each bin is associated with one antenna at the receivingstation and with one tone of the multitone set. The antenna A has fourbins in the memory, one each for Tone 1, Tone 2, Tone 3, and Tone 4.Each bin is further divided into four sub-bins for each of the fourpossible phases, T1, T2, T3, and T4. FIG. 3A shows how the bins andsub-bins in the memory of station Z store the patterns of the white datareceived from station X and the black data received from station Y. Thesignal processor at station Z uses the process of spectral despreading,in accordance with one aspect of the PWAN system, to distinguish thewhite data from the black data. The first user's unique code “1234” isused here as a first spectral despreading code to extract the white dataform all of the signals received by antenna A. The second user's uniquecode “4321” is used here as a second spectral despreading code toextract the black data from all of the signals received by antenna A.After the following discussion of spatial diversity in FIG. 3B, thereader will come to appreciate that there is a similarity in theprocessing of spatially diverse signals and the processing of spectrallydiverse signals, when they are encoded and decoded as has been describedin accordance with one aspect of the PWAN system.

[0133]FIG. 3B is a tutorial diagram illustrating an example of purespatial diversity, showing how a receiver distinguishes two discretemonotone signals from two transmitters that are placed geographicallyfar from one another, in accordance with one aspect of the PWAN system.Transmitting station X receives an input data signal shown as a whitedata signal from a first sender. The first sender is assigned a uniquecode of “1” that is used by the encoder at station X to encode the whitedata signal. In this example, there is only one tone at station X. Theencoder encodes the white data signal onto a single discrete frequencyor tone: Tone 1. The white data signal is copied onto the tone. Themagnitude of a code digit value is converted by the encoder into acorresponding magnitude phase delay in positioning the copy of the whitesignal onto the discrete tone. A phase delay corresponding to a codedigit value of 1 is shown as positioning the copy of the white signal onits discrete tone at the time T1. FIG. 3B also shows transmittingstation Y receiving an input data signal shown as a black data signalfrom a second sender. Station Y is geographically distant from stationX. In this example, the second sender is assigned the same unique codeof “1” at that for the first sender. Thus, the black data signal is alsocopied onto the first discrete tone 1. A phase delay corresponding to acode digit value of 1 is shown as positioning the copy of the blacksignal onto its discrete tone at the time T1. Both stations X and Y haveused the same tone and phase to transmit their respective data. In thisexample, we rely on the diverse geographic locations of stations X and Yto distinguish the signals transmitted from the two stations. This is anexample of spatial diversity. The two signals from station X and stationY are diverse because the phases of their tones will appear arrive atdifferent times at a receiver. This spatial diversity is detected by thereceiving station Z of FIG. 3B, in accordance with one aspect of thePWAN system. FIG. 3B shows the receiving station Z receiving thediscrete monotone signals on its four antennas A, B, C, and D from thestations X and Y. The signals are processed by a signal processorcomputer and stored in a memory. The memory at the receiving station Zis organized so that each bin is associated with one of the fourantennas at the receiving station and with one tone out of a possibleplurality of four tones. If there were four possible tones, then eachantenna A, B, C, and D would have four bins in the memory. As before,each bin is further divided into four sub-bins for each of the fourpossible phases, T1, T2, T3, and T4. FIG. 3B shows how the bins andsub-bins in the memory of station Z store the patterns of the white datareceived on the four antennas A, B, C, and D from station X. FIG. 3Balso shows how the bins and sub-bins in the memory of station Z storethe patterns of the black data received on the four antennas A, B, C,and D from station Y. The signal processor at station Z uses the processof spatial despreading, in accordance with one aspect of the PWANsystem, to distinguish the white data from the black data. The firstuser's unique code “1” and the relative phase delays in the arrival ofthe white data to the four antennas A, B, C, and D is used here to forma first spatial despreading code to extract the white data form all ofthe signals received by antennas A, B, C, and D. The second user'sunique code “1” and the relative phase delays in the arrival of theblack data to the four antennas A, B, C, and D is used here to form asecond spatial despreading code to extract the black data from all ofthe signals received by antenna A. It can be seen by a comparison ofFIG. 3A and FIG. 3B that there is a similarity in the processing ofspatially diverse signals and the processing of spectrally diversesignals, when they are encoded and decoded as has been described inaccordance with one aspect of the PWAN system.

[0134]FIG. 3C is a tutorial diagram illustrating an example of bothspectral and spatial diversity, showing how a receiver distinguishes twodiscrete multitone signals from two transmitters that are placedgeographically far from one another, in accordance with one aspect ofthe PWAN system. Transmitting station X receives an input data signalshown as a white data signal from a first sender. The first sender isassigned a unique code of “3333” that is used by the encoder at stationX to encode the white data signal. In accordance with one aspect of thePWAN system, the encoder uses the discrete multitone spread spectrumprotocol to encode the white data signal onto four discrete frequenciesor tones: Tone 1, Tone 2, Tone 3, and Tone 4. The white data signal iscopied onto each of the four tones. The magnitude of a code digit valueis converted by the encoder into a corresponding magnitude phase delayin positioning the copy of the white signal onto the discrete tone. Theencoder at station X performs spectral encoding in FIG. 3C when itconverts the first sender's unique code of “3333” into positioning fourcopies of the white data at the phases T1 T2, T3, and T4 for Tone 1,Tone 2, Tone 3, and Tone 4, respectively. The station X is shown in FIG.3C as being equidistant from the four receiving antennas A, B, C, and Dof station Z. Thus, the signals arrive at each antenna at the same time.This relative positioning of the sending station X and the receivingstation Z can be viewed as imparting a spatial encoding to the firstuser's white data. The white data is therefore both spatially andspectrally encoded. In accordance with one aspect of the PWAN system,the first sender's unique code “3333” and the relative phase delays inthe arrival of the white data to the four antennas A, B, C, and D isused at station Z to form a first, combined spectral and spatialdespreading code to extract the white data form all of the signalsreceived by antennas A, B, C, and D.

[0135]FIG. 3C shows transmitting station Y located at a geographicposition with respect to station Z so that signals from station Y arereceived at different times by the four antennas A, B, C, and D atstation Z. Station Y receives an input data signal shown as a black datasignal from a second sender. The second sender is assigned a unique codeof “1111” that is used by the encoder at station Y to encode the blackdata signal. In accordance with one aspect of the PWAN system, theencoder uses the discrete multitone spread spectrum protocol tospectrally encode the black data signal onto four discrete frequenciesor tones: Tone 1, Tone 2, Tone 3, and Tone 4. The black data signal iscopied onto each of the four tones. The magnitude of a code digit valueis converted by the encoder into a corresponding magnitude phase delayin positioning the copy of the black signal onto the discrete tone. Theencoder at station Y is shown in FIG. 3C as converting the secondsender's unique code of “1111” into positioning four copies of the blackdata at the phases T1, T1, T1, and T1 for Tone 1, Tone 2, Tone 3, andTone 4, respectively. Since station Y is shown in FIG. 3C as not beingequidistant from the four receiving antennas A, B, C, and D of stationZ, the signals arrive at each antenna different times. This relativepositioning of the sending station Y and the receiving station Z can beviewed as a spatial encoding of the second user's black data. The blackdata is both spatially and spectrally encoded. The second sender'sunique code of “1111” and the relative phase delays in the arrival ofthe black data to the four antennas A, B, C, and D is used at station Zto form a second, combined spectral and spatial despreading code toextract the black data form all of the signals received by antennas A,B, C, and D. Thus, in accordance with one aspect of the PWAN system,signals from both station X and from station Y can be simultaneouslyprocessed by combined spectral and spatial despreading codes todistinguish both spatially diverse signals and spectrally diversesignals, when they are encoded and decoded as has been described inthese examples.

[0136] An exemplary communication system in which the PWAN system may beimplemented is shown in FIG. 3D. In this FIGURE, the various elementsmarked 11 are exemplary fixed remote terminals serving users, while theboxes marked 12 are the base stations associated with certain of thoseremote terminals. It should be noted that in this context the term“fixed” remote terminals applies not only to remote terminals that donot move during use, but may also apply to terminals that are mobile, solong as they are serviced by one base station during a call. Other“fixed” embodiments may include motion between cells during a call andmotion at less than 10 or 5 miles per hour. Additional remote terminalsand base stations are also shown.

[0137] The remotes and base stations are connected by exemplaryairlinks, 13. The base stations may be connected to a “wireless networkcontroller” 14, which then connects to the wider telecommunicationsnetwork, 15. Connections between the base station and the networkcontroller and between the controller and the telephone network, may bewired or wireless.

[0138] An aspect of the PWAN system is centered about the exemplaryairlinks, 13, that connect the base stations and the remotes. Theseairlinks use scarce bandwidth resources and are advantageously operatedin a highly bandwidth efficient mode so as to accommodate a large numberof users.

[0139] The Airlink

[0140] Discrete Multitone Transmission

[0141] The airlink, shown as 13 in FIG. 3D, involves a multitude ofcomplex transmission techniques. The first is an embodiment of multitonetransmission that we call “discrete multitone (DMT)”. In this technique,a signal is transmitted over discrete carrier frequencies, shown in FIG.3E as 21. Specific tones can be assigned to specific users, as shown inthe FIGURE. As discussed above, the tones may be spaced at a frequencyof 1/T, where T is the symbol rate, so that they are “orthogonal”—i.e.,they do not interfere with one another—as in OFDM. Each tone can carrydifferent data, but for the purposes of this discussion, at least someof the various tones assigned to a specific user will be assumed to becarrying redundant information to realize the advantages of frequencydiversity. Such redundant transmission over a range of frequenciesallows recovery of the signal even if some frequencies are subject tosevere interference—a problem of particular interest in the embodimentof this PWAN system that involves fixed remotes. As mentioned above,certain implementations of this signal format enables analysis that canbe effected using fast Fourier transform calculational techniques

DMT-SS

[0142] In an aspect of this PWAN system, bandwidth efficiency isincreased by spreading the signal over a set of weighted tones, witheach user being assigned a specific set of tones and weights. Thistechnique, is depicted in FIG. 3. In this FIGURE, identical data is sentover the four tones identified as 1, 2, 3, and 4. User 1 is to be sent a“+1.” User 2 is to be sent a “+1.” User 3 is to be sent a “+1.” User 4is to be sent a “−1.” The same tones are used to send information to thefour different users by using different “weights” for each user. Theseweights may be viewed as user-specific codes, and we may refer to themas weights, codes or weight-codes. In this heuristic example, theamplitude of a particular tone is obtained by multiplying the data valueby the weight-code value for that combination of user and tone. Forexample, the weight-code of the second user is [1 −1 1 −1], meaning thatfor the second user the amplitude of the first tone is the data valuetimes +1, the value for the second tone is the data value times −1 etc.For example, the value of the second tone for the second user is thedata value, +1, multiplied by the weight-code value for the seconduser's second tone, −1, to yield a −1, as shown in the second positionof the second line. This process is called “spreading” since it effectsthe spreading of the data across the tone set.

[0143] The various tone values are added to obtain the compositespectrum, shown on the last line of the FIG., that is then transmitted.Upon receipt of the spread data, the data is “despread”, i.e., the datato be sent to the various users is obtained, by multiplying thecomposite spectrum by the inverse of a particular user's weight-code.This can be performed simultaneously for all users by using appropriatematrix techniques.

[0144] It is helpful to bear in mind the difference between this“discrete multitone spread spectrum (DMT-SS)” technique and well knownembodiments of the classical spread spectrum technique. In DIRECTSEQUENCE SPREAD SPECTRUM, each data symbol is multiplied by a series ofcode pulses. This spreads the data over a much wider region of thespectrum. In FREQUENCY HOP SPREAD SPECTRUM, the data is transmitted overdifferent regions of the spectrum during different time slots, inaccordance with a pre-defined hopping code. In the DMT-SS used in thisinvention, the signal is modulated by a set of weighted discretefrequencies, not over a continuous broad frequency range, as in directsequence.

[0145] It should be appreciated, that although depicted in this exampleas a set of real numbers, the spreading codes advantageously comprise avector wherein each vector is a complex number.

[0146] Matrix Representation of the “Coding” Process

[0147] Exemplary, high level equipment arrangements used in the“spreading” and “despreading” is represented schematically in FIG. 4. Inthis FIGURE, 41 is the data, D, that is to modulate a DMT-SS signal. At42, the various DMT-SS carriers are encoded as depicted previously inFIG. 3. The mathematical description of the spreading operation is shownin formula 43, where SD is the “spread data,” CM is the “code matrix,”and D is the “data.” The detailed matrix operation is shown in formula44, where 45 is the data vector array representing the data of FIG. 3,46 is the code matrix of that FIGURE, and 47 is the composite spectrumor spread data vector array.

[0148] When the spread data, SD, is received, it is “despread” by meansof the vector operation shown in 60, where SD is the received “spreaddata,” CM⁻¹ is the inverse of the code matrix and DD is the “despreaddata,” that, as required, is reflective of the original data. Thisvector operation is shown in detail in 48, where 49 is the receivedspread data, 50 is the inverse of the code matrix, and 51 is thedespread data. It is important to note for the discussion in the nextsection on the effects of SDMA, that the size of this code matrix isdetermined by the total number of tones that are used.

[0149] As will be discussed below in the section on “Specific Details ofthe Invention,” it is not necessary to use orthogonal codes. In fact, inmost embodiments of this invention the codes are usually only linearlyindependent, and the effects of cross talk between users with differentcodes is treated using a “code nulling” process that resultsautomatically from the practice of one aspect of this invention.

[0150] Use of SDMA in “Coded” Multitone Transmission

[0151] An important aspect of the invention involves the realizationthat the mathematical description of a certain processed spectrallyprocessed signals, such as DMT-SS signals, is analogous to themathematical description of a signal spatially processed by amulti-element adaptive antenna array. Accordingly, the mathematicaldescription of such spectrally processed signals may simultaneouslydescribe spatial processing by a multi-element adaptive antenna array bysimply increasing the size of the DMT-SS matrix to take into account thenumber of antenna elements in the antenna array. The dimensionality ofthe combined “spectral/spatial matrix” that comprises the spreadingweights by which each tone is multiplied, is then equal to the number oftones multiplied by the number of energized antenna elements.

[0152] As noted in the Summary of the Invention, the mathematicalformalism that describes an aspect of this invention treats both codeand antenna aspects of the received signal similarly. The signalprocessing may therefore automatically result not only in codes thathave minimal cross talk with other coded signals, but also in theformation of beams that yield minimal interference from usersilluminated by different beams. These advantages are usually derivedseparately and are known as code nulling and null steering respectively.They will be explained in greater detail below in the section on“Specific Details of the Invention.”

[0153] Channel Response, Equalization, and Signal Extraction

[0154] The discussion to this point has not involved any description ofthe effects of channel response. Distortions due to the channel responsecan be introduced into the formalism by means of a “channel response”matrix, shown in FIG. 5. In this FIGURE, the received data “RD”, shownin 52, is no longer equal to the spread data as in FIG. 4, but is nowdistorted by a channel response “CR.”. The received data is then theproduct of the spread data and the channel response, as shown in 52.This effect is shown in 53, with the exemplar numbers used in FIG. 3. Asshown, the despread data, 54, now do not have the original data values,but rather values that are distorted by the channel response. In orderto correct this distortion, the code despread matrix must include termsthat “equalize” the channel distortion.

[0155] In an embodiment of the invention, this channel response vectoris determined by transmitting a pilot signal and noting its distortionby the channel (“pilot driven equalization”). In another embodiment, theeffect of channel response is “equalized” by simply adaptivelycalculating a “despread matrix” that maximizes the ratio ofsignal-to-noise-and-interference associated with the transmitted data(data driven equalization). The calculated optimum system parameters mayinclude a mathematical representation of the channel response. Thesechannel response parameters may then be used by either the base or theremote to “equalize” the channel distortion. These parameters may beused by the either side of the link because, at least for short periodsof time, the channel is reciprocal in time. Of course, thesecalculations may be equally well done at some more central locationrather than at the remote. What is central to this aspect of theinvention is that certain calculations can be reused. Accordingly,despread weights used on the receive signals can be reused, with onlyminimal modifications, to spread signals on the next transmission—aprocess called retrodirectivity. Additionally, it may be possible toreuse, at the remotes, the weights that are calculated at the basestation.

[0156] Of course, the calculation of optimum system parameters, and theadaptive extraction of the data associated with each user from thecombined signal, is done taking into account all of the system signalsthat are seen by the base and transmitted by the base. Accordingly, theremotes may calculate their own despread weights to take into accountthe interfering signals that they receive, which are not received by thebase station. Simplification may be achieved by using a fixed beampattern for the remotes, rather than calculating beam weights, since theremote knows where the fixed base is and need not reoptimize its beamweights once its beam pattern is fixed.

[0157] Modulation Formats

[0158] Although, to this point, we have shown the signal as either zerosor ones, it is clear that the carrier may be modulated in any one of anumber of signal modulation formats, such as higher order QAM. In suchan exemplary format, the composite spectrum will be as shownheuristically in FIG. 6. In this FIG., the tone sets, as shown in FIG.2, are displayed on the x-axis. The different codes used for the varioustone sets are shown on the z-axis. Finally, the constellationsassociated with QAM modulation are shown on the y-axis as circles. Theparticular constellation point that is energized is represented by aclosed circle. The composite spectrum, obtained by “collapsing” thez-axis, is shown x-z dimension. The blur represents the composite of allof the energized constellations of a particular tone.

[0159] Time Division Duplex

[0160] In an embodiment of the invention, the bandwidth-efficienttransmission techniques used in the invention are combined in a TimeDivision Duplex (TDD) configuration, i.e., a configuration in which thechannel is divided into time slots with uplink and downlink transmissionoccurring alternately in adjoining time slots. A simplistic TDDconfiguration is shown in FIG. 7. As shown in the FIGURE, duringalternate time slots, information is sent uplink (from base station toremote) and then downlink (remote to base station). The guard time isselected to allow for the delay time due to multi-path. All remotes andbase stations may be synchronized so that all remotes transmit at thesame time and then all base stations transmit at the same time. Wellknown GPS techniques may be used for such synchronization.

[0161] As indicated above, the use of TDD, and the assumption of achannel response that varies slowly compared to a TDD period, permitsthe interchangeable use of spread and despread weights, at least duringcontiguous receive and transmit cycles at a given location. Likewise itmay be possible to reuse, at a second location, the larger part of the“spread/despread matrix” calculated at a first location. For example, itmay be possible to use at the remotes the weights—from which each usersinformation can be extracted—that were calculated at a base stationduring a previous TDD period by maximizing thesignal-to-noise-and-interference ratio for the signals received at thebase. In this embodiment, the base sends to its remotes theirappropriate “channel equalized codes” or “weights” using the TDD format.The remotes may perform at least some weight recalculation, but may relyon some of the weight analysis performed at the base station. In thisway the larger part of the calculation may be done at the base stationsor at some other location removed from the remote locations. Thisreduces the cost and complexity of the more numerous remotesconsiderably.

[0162] Of course, to implement this aspect of the invention,optimization parameters must be relatively constant during the timeperiod of one uplink and one downlink time slot. Thereafter newlycalculated optimization parameters may be determined, and sent to theremotes, on a periodic basis.

[0163] Bandwidth on Demand

[0164] As noted above, the invention is particularly well adapted toproviding variable bandwidth on demand. The provision of such additionalbandwidth is effected by simply assigning more tones or codes to therequesting user, or by transmitting in a higher order modulation format.

[0165] Exemplary Details of the Invention

[0166] The Analogy Between DMT-SS and Adaptive Antenna Array Processing

[0167] The aspect of this invention that involves the use of spectralmultiple access techniques that are mathematically analogous to themathematical description of adaptive antenna array signals can be betterunderstood in the context of heuristic FIG. 14. In that FIGURE, 10 isthe mathematical description of Discrete Multitone Spread Spectrum(DMT-SS). In 10, a baseband signal, d(t) is multiplied by a spreadingcode comprising a set of tone frequencies and associated carrierweights, g_(k). It should be appreciated that this is different thanDirect Sequence Spread Spectrum where the baseband signal is multipliedby a PN code, rather than by a set of weighted carriers. The expressionof 10 can be rewritten in block form, as in 20. Here the weightingoperation associated with g_(k) has been separated from the exponentialoperation, that we characterize in the FIGURE as an “inverse frequencychannelizer”, for example an inverse FFT. In a point central to thisaspect of the invention, applicants have recognized that thisrepresentation is analogous to that of an adaptive antenna array—where abaseband signal is multiplied by an aperture vector.

[0168] The DMT-SS despreading operation is shown in heuristic FIG. 15and it also is found by applicants to be analogous to similarexpressions for adaptive antenna array processing. In 10, a widebandsignal x(t) is passed through a bandpass filter, BPF, and thenmultiplied by a despreading code, w(t) nominally the inverse of thespreading code of FIG. 15, to get the original signal, d(t). Bycalculating the despreader weights appropriately, for example tomaximize signal to noise, we can automatically correct for channeldistortion and other interfering signals. In 20, the despreadingoperation is again separated from the exponential coefficients, as inFIG. 14. Here the received signal, x(t) is represented as equal to thesum of an interference term, i(t) and the transmitted signal, s(t)multiplied by a distortion term h(t). From FIG. 14, the transmittedsignal s(t) is equal to g times d(t), giving equation 30 in FIG. 15.Applicants have recognized that this equation is analogous to thatdescribing the output of an adaptive antenna array.

[0169] In accordance with an aspect of this invention, this analogybetween DMT-SS and adaptive antenna array processing leads to thepossibility of combining both spatial and spectral expressions in emathematical expression that can be solved in one unifiedspectral/spatial calculation. This also leads to the further discoveredanalogy between null steering in adaptive antenna arrays and codenulling in CDMA in general, and in DMT-SS in particular. In accordancewith another aspect of the invention, instead of setting the despreadweights to previously estimated spectral spreading and beam steeringweights, we adaptively calculate the despread weights to maximize somegeneral measure of signal quality—either characteristics measureddirectly from the channel or obtained from a blind adaptive operation ora combination of the two.

[0170] Time Division Duplex

[0171] The TDD signaling protocol used in an embodiment of the airlinkis depicted in FIG. 7. It should be noted that two 5 MHz frequency bandsseparated by 80 MHz are depicted in FIG. 7. In one embodiment, the samedata is transmitted over both 5 MHz bands to reduce multi-path fadingeffects. The 80 MHz separation between the two bands ensures that thesame multi-path fade does not interfere with both bands. In addition,the 5 MHz frequency band is divided into four 1 MHz sub-bands, with thelower and upper 500 Hz of each 5 MHz band being designated as guardbands. The four 1 MHz sub-bands in the lower 5 MHz band are matched withcorresponding sub-bands in the upper 5 MHz band. So, for example, thefirst sub-band in the lower 5 MHz band shares is duplicated in the firstsub-band in the upper 5 MHz band, etc.

[0172] In one preferred embodiment, the transmission period from thebase station to the remotes and from the remotes to the base station(T_(symbol)) is approximately equal to 340 microseconds. The guard time(T_(guard)) between transmission and reception is approximately 35microseconds, in one embodiment, while the total revisit time(T_(revisit)) is approximately 750 microseconds. As has been mentioned,and as will be discussed in greater detail below, it is important thatT_(revisit) be less than the amount of time in which significant changesare likely to occur over the air channel, in order to ensure thatessentially the same channel characteristics are observed within anyselected interval of length T_(revisit).

[0173] The guard time, T_(guard), between the forward and reverse burstsmust be sufficiently large to allow significant attenuation ofmulti-path reflections. Base-to-base interference necessitatessufficient guard time between forward and reverse transmission bursts.In an embodiment of the invention four forward bursts are transmittedwithout any intervening reverse bursts and then four reverse bursts aretransmitted. The guard time between the four bursts is much smaller thanthe guard time at the end of the four bursts. This reduces thebase-to-base interference.

[0174] High Level View of the Signal Processing

[0175]FIG. 8 is a signal flow diagram that generally illustrates thesignal processing steps performed in one embodiment of the invention onan audio, video, voice or data signal transmitted over the air interfacebetween the base station and the remotes. As shown in FIG. 8, a signal(that may comprise audio, video, voice or data) is supplied from acommunication link to an input terminal 1010. This signal is thenpacketized in digital format as indicated by a block 1011. The signal ispacketized so that the entire signal can be sent in a single packetduring the transmission time T_(packet). As indicated within block 1012,the packetized signal is subsequently quadrature amplitude modulation(QAM) encoded and error encoded (using, for example, well knownReed-Solomon and/or trellis encoding techniques). Of course, it shouldbe understood that in other advantageous embodiments of the invention,binary phase shift keying (BPSK) or M-ary phase shift keying (MPSK) maybe employed as an alternative modulation technique to QAM.

[0176] The mapper 1012 outputs a complex number representative of ann-bit binary value based upon the mapping scheme. For example, if 16 QAMis used, then the encoder 1012 will output a four-bit binary value, oneof 16 possible values since 2⁴=16. Likewise if 256 QAM is used, then theencoder 1012 will output one of 256 complex values representing aneight-bit binary value since 2⁸=256. The bits that enter the mapper mayhave been forward error correction encoded to protect against channelerrors.

[0177] The encoded signal is then spread over a portion of the frequencyband as indicated within a block 1013. In accordance with an embodimentof the invention, the DMT-SS spreading technique is used to spread theencoded signal over several frequency tones within the total frequencyspectrum. The method used to spread the encoded carrier signal isdescribed in greater detail immediately below with reference to thesignal processing flow diagram of FIG. 9.

[0178] Parallel Data Transmission Using Multitones

[0179] The division and transmission of a signal over a number ofcarriers—Parallel Transmission—is discussed, for example, in a paperentitled “Analysis and Simulation of a Digital Mobile Channel UsingOrthogonal Frequency Division Multiplexing,” by Leonard J. Cimini, Jr.,IEEE Transactions on Communications Vol. Com 33, No. 7, July 1985.Briefly, Parallel Transmission is a signal processing technique thatconverts a serial data stream into a parallel data stream, and modulatesdifferent discrete carrier tones with each of the parallel data streams.

[0180] For example, consider a set of carriers (called a tone set) thatincludes four tones. A serial data stream is then divided into fourparallel data streams by taking every fourth symbol and assigning it toa particular one of the tones. So, for example, the first, fifth, andninth symbols are assigned to the first tone; the second, sixth, andtenth symbols are assigned to the second tone, etc. Accordingly, thefirst tone in the tone set will be set to an amplitude and phasecorresponding to the symbol values output onto the first parallel datastream, the second tone in the tone set will be set to an amplitude andphase corresponding to the symbol values output onto the second paralleldata stream, etc. In a particularly advantageous embodiment of theinvention, the spacing between the tones is carefully selected toprovide orthogonal frequency division multiplexing (OFDM).

[0181] A PN code method of implementing such a modulation scheme, isdepicted in FIG. 9 As indicated above, the data in the parallel datastreams may be the same or different data. The major advantage of thistechnique is that it can be shown that such a processed signal iseffectively the Fourier transform of the original data stream, and thata bank of coherent demodulators is effectively an inverse Fouriertransform. In an aspect of the invention, these technique is used toobtain the calculational advantages of FFT and WFFT processing

[0182] DMT-SS Details

[0183] In an exemplary embodiment of the invention, the total bandwidthallocation for the airlink is 10 MHz. in the range of 1850 to 1990 MHz.The total bandwidth is divided into two 5 MHz bands called the Lower RFband and the Upper RF band. The separation between the lowest frequencyin the Lower RF Band and the lowest frequency in the Upper RF Band (DF)is 80 MHz. The base frequency (f_(base)) for the network is defined asthe lowest frequency of the Lower RF Band.

[0184] The Lower and Upper RF Bands are further subdivided intosub-bands. The first and last 0.5 MHz of each band are designated asguard bands and are hence unused. The remaining 4 MHz in each band isthen subdivided into 4 Sub-bands sequentially numbered from 0 to 3. EachSub-band contains a set of frequencies in the Lower RF Band and anotherset of frequencies in the Upper RF Band. The extension L indicates theset within the Lower RF Band and U indicates the set within the Upper RFBand.

[0185] In one embodiment there are a total of 2560 frequency tonesequally spaced in the 8 MHz of available bandwidth. There are 1280 tonesin each Band, and 640 tones in each Sub-band (320 frequencies in thelower band and 320 frequencies in the upper band). The spacing betweenthe tones (Df) is simply 8 MHz divided by 2560 that translates to 3.125KHz. The tones may be further organized into Tone Sets each with fourtones, and Tone Partitions, each with 20 Tone Sets. Alternatively thetones may be organized into Tone Clusters each with 20 tones, andTraffic Partitions, each with 4 Tone Clusters. A traffic channelrequires at least one traffic partition. Control and access channels maybe interspersed among the traffic channels in the 5 MHz slots. As willbe discussed further below, data is redundant over a tone set.

[0186] The organization of the tones also permits standardization oftone assignments to users so as to permit the contemplated calculationsin an orderly fashion. For example, each user may be assigned onlymultiples of traffic partitions. The division of the total transmissionband into sub-bands also allows for lower sampling rates and lessintensive DSP requirements (since the processed band is spread over asignificantly smaller bandwidth). In addition, the partitions provide aconvenient division for reducing the dimensionality of received vectors.This could be accomplished by combining selected tone set values (i.e.,the corresponding tone set values in each cluster set). Although thisinvolves a reduction in the number of degrees of freedom, such atradeoff can be advantageous in systems wherein the maximum number ofdegrees of freedom are not necessary to accurately decode the data.Thus, by reducing the dimensionality of the tone set vector, theprocessing cost is significantly reduced.

[0187] As indicated above, to ensure that the signals modulated onto theseparate tones do not mutually interfere by overlapping with other tonesthe tones set are spaced at intervals of 1/T, the symbol rate. Ofcourse, some distortion occurs during transmission so that someinterference may occur which may be removed with additional errorcorrecting techniques.

[0188] The Use of DMT-SS

[0189] As noted above, the signals may be initially spread over assignedtones using appropriate codes or weights. These codes may be orthogonalwithin a given spatial cell, and may be randomly assigned to the sametone bins within adjacent spatial cells. Thus, spreading codes may bereused in adjacent spatial cells and also may have a random correlationbetween adjacent spatial cells. Although the initial code assignmentsmade by the base station may be orthogonal, it will be understood thatin response to the weight adjustments made during adaptive equalization,the spreading codes will typically evolve, or adapt, to non-orthogonalcodes after the communications network has been active for some time. Aswill be discussed in greater detail below, the criterion for thespreading codes used within a given spatial cell is advantageouslylinear independence rather than orthogonality. The random correlation ofspreading codes in adjacent spatial cells is compensated for by means ofan automatically implemented code nulling technique that nulls outcorrelated portions of the transmitted signals using linear weighting.

[0190] Once the spreading codes associated with the DMT-SS modulationtechnique have been assigned to the encoded data signals, as representedby the block 1013 of FIG. 8, the processed signals are linearly summed,as indicated by a summing block 1025. A similar signal processingprocedure is used on other incoming signals as indicated by the blocks1021-1023, that correspond to the blocks 1011-1013. These signals aresummed within the summer 1025 and assigned to carrier frequencies withinthe 5 MHz sub-bands shown in FIG. 7, as indicated by a block 1030.

[0191] As noted above, during the course of adaptive equalization, thecodes typically become non-orthogonal in order to maximize the SINRthroughout the overall communications network 100. However, in order toretain the maximum number of degrees of freedom throughout thecommunications system 100, it is preferable to maintain linearindependence of the complex spreading vectors throughout a given spatialcell. Linearly independent complex vectors are those that cannot beexpressed as a sum or scalar multiple of any combination of the othercomplex vectors in the system Thus, by preserving linear independenceamong the spreading codes, a matrix set of linear equations can bederived that allows each of the system variables (i.e., data symbols) tobe uniquely decoded. Insofar as the spreading codes become more linearlydependent, the ability to discriminate amongst data symbols becomes moredifficult. However, in some applications, band pass filter values areestablished in the beginning and thereafter the system must operatewithin those constraints.

[0192] The spread signals are linearly added on a carrier-by-carrierbasis to obtain the overall DMT-SS waveform. In order to despread thissignal, the received signal is detected and converted into matrix form.The received vector is multiplied by a scaling factor (that isproportional to the number of bits in the spreading code), and a matrixcomprised of the spreading codes. The resultant vector provides thedespread data symbols as an output. From this example, it can be inducedthat as many data bits can be distinctly despread as there are bits inthe spreading codes, so long as the spreading codes remain linearlyindependent

[0193] Returning to the spreading of the data, once the encoded,spread-spectrum signals have been assigned to the frequency carrierbands, the signal may be transformed from the discrete frequency domainto the analog time domain using an inverse fast Fourier transform (EFFT)and an analog to digital converter. By using an IFFT and an FFT toprovide for OFDM, multiple modulators are not required, as is well knownin the art. This is because the calculations relating to the DMT-SSmodulation technique are less intensive in the frequency domain than inthe time domain. For this reason, the bulk of the signal processing ispreferably performed in the frequency domain (with the exception of themodem operations of, for example, encryption, filtering, etc.) and istransformed to the time domain as one of the last steps beforetransmission.

[0194] Depending upon bandwidth considerations, signals from the sameuser are assigned one or more spreading codes and one or more trafficpartitions. The assignment of different spreading codes and additionaltraffic partitions to provide additional bandwidth for a requesting userunit (a unit that communicates via one of the remotes) is particularlyelegant (in comparison with bandwidth allocation using TDMA) from animplementation standpoint. This is because the allocation of newspreading codes and tone sets is mathematically simple and merelyrequires a numerical change to the despreading vector (for thereassignment of a new tone set) or an increase or decrease of thebandwidth of a bandpass filter on the receiving side (for thereassignment of a new tone set).

[0195] Advantages Associated with DMT-SS

[0196] The use of DMT-SS is highly advantageous in the system of thepresent invention For example, the use of DMT-SS allows the channelcharacteristics to be evaluated at discrete points that can be exactlyrepresented in matrix form as a complex vector. Thus, because selectedtones within each tone set can be designated as pilots distributedthroughout the frequency band, a simple evaluation of a finite number ofcomplex values results in an accurate channel estimation. Furthermore,theoretically, the channel distortion can be exactly compensated at thediscrete tone frequencies by a simple complex conjugate multiplication.That is, since discrete tones are used, it is not necessary to know theentire channel response between the tones since the channel only affectsoperations at the exact points of the tone set frequencies. If thechannel is defined at these discrete points, the received tones needonly be multiplied by the appropriate complex, amplitude and phase toequalize the channel. This means that exact equalization is accomplishedby a simple complex multiplication. This channel equalizationcalculation may be subsumed in the calculation of despread/spreadweights that improve or optimize characteristics of the signal such asthe signal to noise and interference ratio.

[0197] Also, the use of DMT-SS ensures that the equalization of antennaarray time dispersion is very simple. In multiple element antennaarrays, a time delay is observed between receptions of a waveform by thespatially separated sensors when the wave impinges on the array. In avery wide band system, this delay creates dispersion. However, by usingDMT-SS, the dispersion can be represented by discrete values of ascaleable vector since the response is only evaluated at discrete pointsof the frequency.

[0198] Furthermore, each user on the system could operate with adifferent QAM (or other M-ary) constellation size. This is because thesymbols are not spread over the entire bandwidth as in direct sequencespread spectrum. Rather, in DMT-SS the symbols are spread over frequencybins of various sizes so that each user can have the optimum size QAMconstellation (i.e., the highest order allowable in a given SINR). Thisincreases the overall system capacity since the system is not restrictedto the lowest common denominator (i.e., the QAM or M-ary constellationsize at which all channels can operate). In addition, at lowerconstellation sizes a lower signal-to-noise ratio is required todemodulate the signal, and this lower signal-to-noise ratio requirementcan be used to extend the range of the base station that providesadditional system flexibility. Finally, the packet size could also bevaried in conjunction with the QAM constellation size to manageinterference levels.

[0199] The use of DMT-SS modulation also provides several unexpectedadvantages when used in combination with certain communicationtechnologies First of all, since DMT-SS spreading allows for flexiblespreading bandwidths and gain factors (i.e., a given signal can bespread over as much bandwidth as desired), it is particularlyadvantageous for exploiting the spectral diversity of the channel. Thatis, since the channel has certain bands with better response than otherbands, signals can be selectively spread over the more desirable bands.

[0200] In addition, DMT-SS also allows for the use of code-nulling togreatly improve the reuse capacity of the communication link beyond thereuse capacity of conventional CDMA. Since DMT-SS is used instead ofdirect sequence or frequency hopping, selected portions of the spreadingcode can be nulled within the despreader. Thus, only those portions ofthe spreading code which are not common with the interfering spreadingcodes will be despread. Furthermore, DMT-SS is particularly advantageouswhen implemented within a variable bandwidth system since the allocationof bandwidth is highly flexible in such a system, and can be implementedby the appropriate assignment of additional tones to the requestinguser. In summary, DMT-SS provides a solution that nulls the interferingsignals.

[0201] Finally, DMT-SS is advantageous as applied to a multi-elementantenna array system where matrix calculations comprise the bulk of theprocessing operations. As is well known in the art, as thedimensionality of a matrix grows, the calculation operations necessaryto invert the covariant matrix increases as the cube of the matrixdimensionality. Thus, the processing power increases as the cube of thematrix dimensionality and, consequently, so does the cost of theprocessing circuitry. Thus, in order to avoid skyrocketing costs, it isadvantageous to limit the dimensionality of the matrices used to performthe spreading and despreading calculations. Since in a multi-elementantenna array system it is sometimes desirable to change the number ofantenna sensor elements to enhance the beam forming capability of thesystem, such a system would normally incur an increase in matrixdimensionality (since each sensor corresponds to an element in thematrix). However, in a DMT-SS system, if sensors are added to theantenna array, the dimensionality of the matrix can be preserved byreducing the number of tones in each tone set.

[0202] This preservation of matrix dimensionality is possible becausethe mathematical formalism used when performing amplitude and phaseweighting of the signal on each of the sensors is substantially similarto the formalism used when performing amplitude and phase weighting ofeach of the tones in a tone set. Thus, an analogy exists between themultiple sensors in an antenna array and the multiple tones in a toneset. Consequently, the same matrix can be used to determine weights forboth sensor elements and tones, so that if the number of sensor elementsincreases, the number of tones can be decreased to compensate (i.e.,preserve the same matrix dimensionality), and vice versa. Furthermore,essentially the same SINR is preserved in such a system since thedegrees of freedom lost in the number of tones is regained in the numberof beams. In contrast, direct sequence spread spectrum could not changethe number of tones as beams are added since there are no tones to addor subtract. Thus, the cost of such a system would increase enormouslyrelative to the cost of the system of the present invention as capacityis increased. Specifically, the cost of the present invention increasesapproximately proportionally with the capacity, while the cost ofanother system using, for example, direct sequence spread spectrum,increases as the cube of the capacity.

[0203] Once the signal has been DMT-SS modulated, the signal is outputto the antenna for transmission. Control circuitry within the basestation keeps track of the location of the user terminals incommunication with the base station, so that the appropriate signals aredirected to the appropriate user units (i.e., by means of antennabeam-forming discussed below).

[0204] Beam Forming

[0205] In accordance with one aspect of the present invention, adaptiveantenna arrays are used in conjunction with a beam forming algorithm toachieve spatial diversity within each spatial cell and implement SDMA.That is, signals output by the antennas are directionally formed byselectively energizing different antenna sensors with different signalgains so that remote terminals in one portion of a spatial cell are ableto communicate with the base station while other remote terminals in adifferent portion of the spatial cell may communicate with the same basestation, even if they are using the same tone set and code. It should beunderstood that in the fixed implementation of the current invention,i.e., where the remote access terminals do not move substantially duringcommunication with the base station, usually staying within a spatialcell during communication, the beam forming algorithm used in theairlink need not account for mobile remote units leaving and enteringthe spatial cell. In one advantageous embodiment, each spatial cell ispartitioned into four sectors where each sector transmits and receivesover one of the four sub-band pairs.

[0206] As set forth above, the beam forming method of the presentinvention, like the use of codes, should not be conceived as separatefrom the overall adaptive equalization method of the present invention.Rather, the method used to selectively energize the antenna sensors(during transmission) or selectively weight the signals received on thedifferent sensor elements (during reception) is subsumed into theoverall method used to maximize SINR The relation of the beam formingmethod to the overall maximization of SINR method will be described ingreater detail below.

[0207] Code-Nulling

[0208] The use of spread-spectrum technology (particularly DMT-SS) anddirectional antennas within the preferred airlink of the presentinvention allows for several error cancellation benefits, includingeffects that are analogous to code nulling and null steering, by meansof linear weighting in code and space.

[0209] Code-nulling is used to discriminate between non-orthogonalsignals emanating from adjacent spatial cells. Again, the code-nullingmethod should be understood in the context of the maximization of SINRmethod of the present invention. That is, the code-nulling method shouldbe considered as the portion of the method that maximizes SINR withrespect to the code domain. This way of understanding the code-nullingmethod will be described in further detail with respect to FIG. 10.

[0210] It should be understood that if signals generated within the samespatial cell or beam all have orthogonal spreading codes, code-nullingis typically not necessary since the orthogonality is sufficient toensure that there is no cross modulation. However, as mentioned above,the spreading codes used within a particular spatial cell may not beorthogonal, although they are preferably linearly independent.Furthermore, the transceivers within the neighboring spatial cells mayemploy spreading codes that have a random correlation with the spreadingcodes used in the local spatial cell.

[0211] By adjusting the spreading weights associated with eachcommunications channel the base station is able to cross-correlate thesesignals on the same tone set to subtract out interference due to“neighboring” signals. In one embodiment, the base station has thespreading codes used to spread different signals assigned to the sametone set, so that this information can be used to initially calculatethe appropriate weights for nulling out interference from other codes.

[0212] As discussed above, when the spreading codes used to spreaddistinct data signals are orthogonal, the spread data can be preciselyrecovered during despreading. However, when the spreading codes are notorthogonal (as is the case with spreading codes that are used inneighboring spatial cells), cross modulation may result so that the datasignals are not able to be precisely distinguished by simple despreading(i.e., despreading without code-nulling).

[0213] In order to compensate for this phenomenon, code-nulling weightsused in the despreader are multiplied by the received signal vector. Bynulling out the cross modulation present in the received signal, theappropriate values of the data bits are output by the receiver. As longas the complex spreading weights are linearly independent, and the SNRis sufficiently high, the exact symbol values can be discriminated bythis method. It will be appreciated that the code-nulling procedureabove is inherently implemented during derivation of the overall weightsthat maximize the SINR.

[0214] Null-Steering,

[0215] In addition to code-nulling, an exemplary directional antennashown in FIGS. 11 and 12 with no spectral spreading, forms signalsincluding null regions (i.e., regions where the antenna attenuatesincoming signals or where there is a very low antenna gain) These nullregions can be formed in a pattern so that the nulls are directedtowards known interferers (e.g., interfering signal sources orinterfering multi-path reflectors). In this manner, interfering signalsare de-emphasized in the spatial domain. As will be discussed in greaterdetail below, the use of null-steering in conjunction with code-nullingis highly advantageous.

[0216] In accordance with one aspect of the present invention,significant processing time and sophistication can be saved sincesignificant similarity exists between the methods for performingnull-steering and code-nulling. Specifically, the mathematical formalismused to achieve null-steering is analogous to the formalism used toachieve code-nulling. According to this analogy, just as the tones in atone set are multiplied by complex weights to alter the amplitude andphase of the tones, so are the gain and relative phase of signals outputand received by the antenna elements altered by a set of multiplicativeweights. This multiplication by complex weights can be expressed in amatrix form for both code nulling—a spectral concept—and null steering—aspatial concept. Thus, the calculations performed in the spectral codedomain correspond formally to the calculations performed in the spatialdomain. Consequently, null steering can be performed in a system usingcode-nulling simply by adding an extra dimension to the matrices usedfor calculating the complex weights and multiplying the signals by theseweights.

[0217]FIG. 10 generally depicts how weights calculated in both the codeand spatial domain are used to maximize the SINR. It should be notedthat FIG. 10 is primarily a conceptual representation and is not meantto convey the actual processing steps that occur in the method ofmaximizing SINR. As shown in FIG. 10, a three dimensional graph plotsthe relationship among code, space, and SINR. Specifically, the code andspatial domains are shown in one plane, while the SINR is plottedperpendicular to the plane defined by the code and spatial domains. TheSINR is plotted on a scale of 0 to 1 where a value of 0 indicates thatthe signal consists entirely of noise and interference while a value of1 indicates that the signal consists entirely of the signal of interest.

[0218] The code domain axis of the graph represents the variousweighting values that can be applied to each of the tones, while thespatial domain axis of the graph represents the weighting values thatcan be applied to each of the antenna elements. As can be seen from theplot of FIG. 10, certain weights applied in the correct combination ofcode and spatial values result in SINR values near 1 so that optimalsignal detection is achieved by calculating the code and spatial weightsthat converge to the “peaks” depicted in FIG. 10. The method of alteringthe code and spatial domain weights so that convergence to the peak STNRis achieved is described in greater detail below with reference to themethod of maximizing SINR section. The invention combines spatial andspectral spreading and despreading to optimally remove interference fromthe received signals.

[0219] Returning to the null steering procedure that forms a portion ofthe method for calculating weights in the spatial domain, the nullsteering method, illustrated schematically in FIG. 13, provides forincreased user capacity for each base station. As depicted in FIG. 13, afirst beam, “beam A,” is directed by the antenna 120 using beam-formingtechniques, over a particular spatial region (i.e., the signal strengthis significant in the depicted region enclosed by solid lines). A secondbeam, “beam B,” is directed by the antenna 120 over a different spatialregion (enclosed by the dashed line in FIG. 13). Both signals includesidebands, that normally would generate interference within the adjacentsignal space, and null regions between the main beam and the sidebands.Of course, it will be appreciated that more complicated beam patternsmay be employed having several sidebands and null regions.

[0220] In accordance with one embodiment of the invention, the nullregions of beams A and B are positioned in the direction of each of theinterfering transceivers (e.g., transceivers operating on the same toneset and/or code as the intended transceiver). Thus, as depicted in FIG.13, while beam A is directed towards remote A (since remote A is theintended receiver) the null of beam A is directed towards remote B(since remote B is an interferer). Similarly, beam B is directed towardsremote B (since remote B is the intended receiver) while the null ofbeam B is directed towards remote A (since remote A is an interferer). Asimilar weighting scheme is observed when the remotes are transmittingand the base station is receiving. The same null-steering principle alsomay be applied to reduce the interference due to neighboring basestations.

[0221] It should be noted here that multi-path reflectors may also betreated as interfering signal sources so that null regions can bepositioned to null out signals from these reflectors. However, in oneembodiment, if the reflectors are not significantly time varying, thereflected interferers are not nulled. Rather the reflected signals areadvantageously phase shifted to provide constructive interference sothat the SINR is increased.

[0222] The null resolution (i.e., the closeness in degrees of the nulls)which the antenna arrays are capable of providing is dependent uponseveral factors. Two main factors are the spacing of the antenna sensorelements and the S/N ratio of the incoming signal. For instance, if theaperture size is sufficiently large (e.g., if the sensor elements aresufficiently far apart) then a better null resolution will result. Also,if the S/N ratio of the received signal of interest is high enough, thenthe signal of interest could actually be placed partially within a null(so that some gain of the signal is lost, but the overall ratio betweenthe gain null on the interferer and the gain null on the signal ofinterest allows for effective cancellation of the interferer anddetection of the signal of interest). For example, if 15 dB of gain isnecessary to close the link for a given channel, and the S/N ratio ofthe signal of interest is 30 dB, while the S/N ratio of the interfereris 60 dB, then if a null of −70 dB is placed on the interferer, whilethe signal of interest is in the same null at about −15 dB, then theinterferer will have a net −10 dB gain and the signal of interest willhave a net 15 dB gain so that the interferer is canceled and the link isclosed. Thus, a higher S/N ratio allows the nulls to be placed closer tothe signals of interest so that a higher null resolution is achieved. Itshould be noted here that, in accordance with one advantageousembodiment of the invention, the depth of a given null is proportionalto the strength of the interferer that is to be canceled. In addition,due to the frequency diversity provided by the system, nulls canactually be collocated if the steering vectors (associated with the codeweights) of two interfering remotes are sufficiently distinct to providethe necessary processing gain to close the communications link.

[0223] In an alternate embodiment, the remote terminals also includedirectional antennas in one preferred embodiment so that the remoteterminals are also capable of null steering. FIG. 16 is a graph plottingantenna gain (measured in decibels) versus direction (measured indegrees). A number of base stations are represented in FIG. 16 bycrosses, while other remotes (having non-orthogonal codes) arerepresented by small circles.

[0224] In the worst case scenario, the remote is located equidistantfrom three base stations (i.e., on a vertex of a hexagonal spatialcell). This case is represented in FIG. 16 by the presence of threecrosses that transmit with substantially equivalent signal strength.These base stations are shown at approximately 0, 90°, and −90° from thezero direction of the remote antenna.

[0225] Normally, each of the base stations would be received at the samelevel (i.e., at −85 dB) so that substantial interference would resultbetween the three base stations when received at the remote. However,due to the beam forming weights applied by the directional antenna ofthe remote, the interfering base stations (i.e., the stations at ±90°)are attenuated by approximately 50 dB (i.e., 120 dB minus 70 dB)relative to the intended base station (i.e., the base station at 0°).Thus, due to the fact that the beam from the receiving remote antenna isformed to have maximum gain at the intended base station, and to haveminimum gain (nulls) at the strongest interfering base stations, theremote terminals are able to more easily discriminate between the signalof interest and interfering signals. That is, by means of beam formingand null-steering employed at the remote terminals a much highersignal-to-interference plus noise ratio (SINR) can be obtained in muchthe same manner as with the base stations.

[0226] It should be noted here that the remote terminals may also employcode nulling. In an alternate embodiment, initial code nulling weightsare calculated within the base station and transmitted to the remoteterminals. The remote terminals subsequently adapt the transmittedweights to maximize the SINR as required by the particular interferenceenvironment of each remote. By calculating the initial weights andsending these to the remote terminals, much of the intensivecalculations need not be performed within the remotes. Thus, the remoteterminals can be made more cost effectively.

[0227] In one aspect of the invention—referred to as“retrodirectivity”—the base stations adapt the spreading and despreadingweights used within the base stations for transmitting and receivingsignals in order to maximize the overall SINR within the communicationsnetwork 100. In an alternate embodiment, this may be performed, forexample, by monitoring the average bit error rate (BER) throughout thecommunication network 100 and modifying the spreading weights at each ofthe base stations, as well as each of the remote terminals under thecontrol of their respective base stations, to decrease the BER.

[0228] Despread Weight Adaptation Algorithm

[0229] In one embodiment of the present invention, during the trafficestablishment phase, a series of pilot tones having known amplitudes andphases, are transmitted over the entire frequency spectrum. The pilottones are at a known level (e.g., 0 dB), and are spaced apart byapproximately 30 KHz to provide an accurate representation of thechannel response (i.e., the amplitude and phase distortion introduced bythe communication channel characteristics) over the entire transmissionband. To compensate for the channel distortion, a complex inverse(having an amplitude component and a phase component) of the channelresponse is calculated and multiplied by the incoming signals. Thisinitializes the weights during the traffic establishment phase.

[0230] In certain cases, where the channel induced fade is too deep toprovide an adequate signal-to-noise ratio, the tone clusters where thesedeep nulls occur are excised (i.e., discarded so as to not factor intothe signal during despreading).

[0231] Since the channel response varies over time, the set of complexconjugate compensation weights are periodically recalculated to insurean accurate channel estimation.

[0232] Another method of channel equalization involves equalizing thechannel effects (due, for example, to noise and known interferers) bydata directed methods That is, rather than transmitting a known trainingsignal (such as a set of pilot tones), weights are applied to thereceived signal so as to detect a selected property of the data signal.For example, if a PSK modulation technique is used on the data, aconstant power modulus is expected in the received signal. Alternately,in a QAM signal, the data will be detected in an amplitude-phase signalconstellation plane to have substantially concentric rings. Thus, if thechannel is equalized in such a manner as to obtain the desired signalcharacteristics, there is a high probability that the transmittedsymbols will be accurately decoded at the receiver. This generaltechniques is referred to as a property restoral technique. In oneembodiment of the invention the property that is restored is the finitealphabet of the QAM or M-PSK symbol.

[0233] Of course, it will be appreciated by those skilled in the artthat although the channel equalization method used in accordance withthe invention is conceptually separable from other signal weighting anddecoding methods of the present invention (discussed below), the channelequalization method may implicitly include multiple cancellation anddespreading methods. Therefore, the adaptive channel equalization methodof the present invention used to maximize the SINR should not beconsidered as a separate method from the additional methods describedbelow that refer to interference cancellation and signal despreading anddecoding methods. Rather, the adaptive channel equalization method ofthe present invention should be understood to encompass a plurality ofthe below described methods.

[0234] Reciprocity and Retrodirectivity

[0235] TDD is particularly advantageous in the practice of thisinvention since with the use of TDD the linear weighting coefficientsused to compensate for channel interference during transmission andreception of the encoded signals need not be re-calculated within astation. The short time duration between transmission and reception bythe base station, the fact that the transmission and reception occurs inthe same frequency band and only slightly separated in time (TDD), andthe fact that the remote access terminals are stationary with respect tothe base stations assures that the channel is approximately reciprocal.That is, the properties of the air channel between the base and theremote terminals (i.e., those properties that introduce distortion inthe transmitted signal) are substantially the same for both receptionand transmission. Thus, substantially the same weights can be used at astation for both despreading a signal at reception and for spreading asignal at transmission. In accordance with this retrodirectivityprinciple, the base station can perform most of the computation fortransmission spreading weights when it computes the despreading weightson reception. The transmission spreading weights are merely scalarmultiples of the reception despreading weights Similarly, in accordancewith this retrodirectivity principle, the remote station can performmost of the computation for its transmission spreading weights when itcomputes its despreading weights on reception.

[0236] In an alternate embodiment of the invention, the base station cantransmit the weights to the remote stations to be used in the nextreception at the remote station. In this manner, processing is reducedwithin the remote stations since a large portion of the intensivecalculations are performed solely within the base station. Thus, insteadof being prohibitively sophisticated, the remote terminals can be madeat a suitable size and at a reasonable expense.

[0237] Because each remote terminal stands in a different spatialrelation to the other remotes and bases within the communicationsnetwork, each remote terminal advantageously uses equalization weightsthat are individually set to maximize the SINR of signals transmitted toand received from the base station to which the remote is assigned. Thismay be accomplished in different ways. For instance, the base stationmay pre-emphasize the signals sent to the remote by a calculated set ofweights. Since the pre-emphasis approximately compensates for channeldistortion, the remote need not perform weight adjustment calculationsthat are as intensive as those calculated by the base station. Thus, theremotes need not include prohibitively sophisticated processingcircuitry to implement this feature of the invention.

[0238] In one aspect of the invention optimum transmit weights arecalculated based on the signals received at the base station. This iscalled retrodirectivity. When retrodirective adaptive equalization isused to determine the set of weights used in both reception andtransmission, network-wide retrodirective adaptive equalization isaccomplished. Thus, the channel characteristics throughout the entiresystem are accounted for in accordance with this aspect of the presentinvention.

[0239] Of course, as with other aspects of the invention, it will beappreciated that the reciprocity and system-wide retrodirective aspectsof the present invention may also have application in a mobileenvironment. Specifically, if the time duration between transmission andreception in the TDD system is made sufficiently small, the channel mayalso be reciprocal for mobile transceivers so that the same principlesset forth above apply in the mobile environment.

[0240] Zone Control In a particularly preferred embodiment of theinvention, a zone controller could be used to minimize the risk ofinterference between remote terminals that are near one another inadjacent spatial cells. According to this aspect of the invention, thezone controller is informed of the locations of each of the remotes andbase stations within an assigned zone. Those remote terminals that arelikely to interfere are assigned different codes and tone sets tominimize the risk of interference.

[0241] Bandwidth-on-Demand

[0242] In accordance with one aspect of the present invention,bi-directional communication is established between multiple remote userunits and a telephone network via the high-bandwidth base station on auser-by-user basis. Each remote user unit upon activation, initiatescommunication with the high-bandwidth base station by indicating to oneof the remote terminals, included within the remote user unit, theamount of bandwidth desired by the remote user unit. The remoteterminals communicate with the base station via a control channelthrough the air (i.e., the airlink). The high-bandwidth base stationthen sends information concerning the requested bandwidth to a centralbandwidth controller, shown in FIG. 17, that determines whether or notthe requested bandwidth can be allocated to the requesting remote userunit. In this manner, bandwidth is dynamically allocated based upon thetype of user unit and the kind of data that is to be transmitted. Asindicated above varying amounts of bandwidth may be assigned byallocating additional tone sets to the requesting user.

[0243] III A Specific Embodiment of the Invention

[0244] The following description is a specific embodiment of theinvention that includes many aspects of the description provided above.However, it should not be interpreted to limited the scope of theinvention in any way

[0245] Frequency Definitions

[0246] The total bandwidth allocation for the airlink of this specificembodiment of the invention is 10 MHz in the range of 1850 to 1990 MHz.The total bandwidth is divided into two 5 MHz bands called the lower RFband and the upper RF band. The separation between the lowest frequencyin the lower RF band and the lowest frequency in the upper RF band (DF)is 80 MHz. The base frequency (f_(base)) for this embodiment is definedas the lowest frequency of the lower RF band. FIG. 18 shows the possibleoperational bands for this embodiment.

[0247] The lower and upper RF bands are further subdivided intosub-bands as shown in FIG. 19. The first and last 0.5 MHz of each RFband are designated as guard bands and are hence unused. The remaining 4MHz in each RF band is subdivided into four sub-bands sequentiallynumbered from 0 to 3. Furthermore, the suffix “A” indicates a sub-bandwithin the lower RF band and “B” indicates a sub-band within the upperRF band. The sub-bands are paired with each sub-band pair containing onesub-band from the lower RF band and another from the upper RF band.

[0248] There are a total of 2560 tones (carriers) equally spaced in the8 MHz of available bandwidth. There are 1280 tones in each band. Thespacing between the tones (Df) is thus MHz divided by 1280, or 3.125KHz.

[0249] The total set of tones are numbered consecutively form 0 to 2559starting from the lowest frequency tone. T_(i) is the frequency of theith tone:

T _(i) =f _(base) +f _(guard) +Df/2+(i)(Df)

for 0≦i ≦1279

T _(i) =f _(base) +DF+f _(guard) +Df/2+(i)(Df)

for 1280<i<2559

[0250] where f_(base) is the base frequency defined in Table 2.3,f_(guard) is 0.5 MHz, Df is 3.125 KHz, and DF is 80 MHz. Equivalently,the relationship may be expressed as:

T _(i) =f _(base)+500+(i+1/2)(3.125 KHz)

for 0≦i ≦1279

T _(i) =f _(base)+80500+(i+1/2)(3.125 KHz)

for 1280≦i ≦2559

[0251] Each sub-band pair contains 640 tones (320 frequencies in thelower band, and 320 in the upper band). The mapping of tones to eachsub-band is shown in FIG. 20. The set of 2560 tones is the tone space.The tones in the tone space are used to transmit two types of data:traffic data and overhead data. The tones used for transmission oftraffic are the traffic tones, and the rest are the overhead tones.

[0252] Traffic Tones

[0253] The traffic tones are divided into 32 traffic partitions denotedby P₀ to P₃₁. (In this embodiment a traffic channel requires at leastone traffic partition.) Each traffic partition contains 72 tones asshown in FIG. 21. Tone mapping into the ith traffic partition (P_(i)) isshown in Table 2.5.

[0254] Overhead Tones

[0255] The overhead tones are used for the following channels:

[0256] Forward channels:

[0257] The Common Link Channel (CLC) used by the base to transmitcontrol information to the Remote Units;

[0258] The Broadcast Channel (BRC) used to transmit broadcastinformation from the Base to all Remote Units; and

[0259] The Remote Unit Synchronization Channel (RSC) used by the base,for example, to transmit pilot signals, frame synchronizationinformation.

[0260] Reverse channels:

[0261] The Common Access Channels (CACs) is used to transmit messagesfrom the Remote Unit to the Base; and

[0262] The Delay Compensation Channel (DCC) used to adjust a RemoteUnits TDD timing.

[0263] For each sub-band pair, there is one grouping of tones assignedto each channel. These groups of tones are referred to by the name oftheir channels and their sub-band pair index (0, 1, 2, or 3). Forinstance, the CLC channel in sub-band pair 2 is denoted by CLC₂.

[0264] There are two different CACs in each sub-band pair: CAC_(i, 0),and CAC_(i, 1), where i is the sub-band pair index. The two channels maybe used as either solicited (SCAC) or unsolicited (UCAC). The allocationof tones to each of these channels for the ith sub-band pair is shown inFIG. 23. Indices are provided for all tones within a given channel. Theabsolute tone index within the tone space can be determined by therelationships shown in FIG. 23. For instance:

[0265] For the forward channel, the 13th tone in the CLC channel insub-band pair 2, is denoted by CLC₂(13) and its absolute tone index is:

CLC ₂(13)=T _(320.2+1460) =T ₂₁₀₀

[0266] For the reverse channel, the 13th tone in the first CAC channelin sub-band pair 2, is denoted by CAC_(2, 0)(13) and its absolute toneindex is the same as above. FIG. 24 provides a pictorial representationof the division of the tone spaces into different tone groupings

[0267] Time Definitions

[0268] TDD is used by Base and the Remote Unit to transmit data andcontrol information in both directions over the same frequency channel.Transmission from the Base to the Remote Unit are called forwardtransmissions, and from the Remote Unit to the Base are called reversetransmissions.

[0269] As shown in FIG. 25, the duration of a forward transmission isT_(forward), and the duration of a reverse transmission is T_(reverse).The time between recurrent transmissions from either the Remote Unit orthe Base is TTD, the TDD period. A guard period of duration T_(f-guard)is inserted between the forward and reverse transmissions, and a guardperiod of duration T_(r-guard) is inserted between the reverse andforward transmissions.

[0270] As shown in FIG. 26, in every TDD period, there are fourconsecutive transmission bursts in each direction. Data is transmittedin each burst using multiple tones. The burst duration is T_(burst). Aguard period of duration T_(b-guard) is inserted between each burst.FIG. 27 shows the values of the TDD parameters.

[0271] In addition to synchronizing and conforming to the TDD structuredefined in the last section, both the Base and the Remote Unit mustsynchronize to the framing structure. The framing structure is shown inFIG. 28. The smallest unit of time shown in this figure is a TDD period.Two TDD periods make a subframe, eight subframes make a frame, and 32frames make a superframe.

[0272] Frame synchronization is performed at the superframe level. Theframe and subframe boundaries are determined from the superframeboundary.

[0273] In this embodiment we could potentially reuse all availablefrequencies in every spatial cell. However, initially a reuse factor of2 is used. Each Remote Unit is assigned to a Sub-band Pair depending onits location within the spatial cell and the traffic loading of theSub-band Pair. As shown in FIG. 29 each Remote Unit may be assigned twoof the four sub-band pairs depending on its location. For example, anRemote Unit in the north-eastern part of the spatial cell in FIG. 29 canbe assigned Sub-band Pair 0 or Sub-band Pair 2. Of course this reusestrategy reduces capacity to half of the maximum potential capacity. Thesame Sub-band Pair assignment is used in all spatial cells as shown inFIG. 30.

[0274] Forward Channel Format

[0275] The physical layer has three possible implementations based onthe desired range (or quality) of transmission. The physical layermanages the trade-offs between bandwidth efficiency (bits/symbol) andtransmission coverage by providing three modes of operation:

[0276] High capacity mode (short range): 3 bits/symbol

[0277] Medium capacity mode (medium range): 2 bits/symbol

[0278] Low capacity mode (long range): 1 bits/symbol

[0279] Each mode employs different details in the coded modulationscheme and, hence, somewhat different formats. Nevertheless, there isabundant symmetry, redundancy, and common elements for the three modes.

[0280] High Capacity Mode

[0281] In high capacity mode, one traffic partition is used in onetraffic channel. In medium and low capacity modes, two and three trafficpartitions are used, respectively. The Base transmits information tomultiple Remote Units in its spatial cell. This section describes thetransmission formats for a 64 kbits/sec traffic channel, together with a4 kbps Link Control Channel (LCC) from the Base to a single Remote Unit.The block diagram for the upper physical layer of the Base transmitterfor high capacity mode is shown in FIG. 31, which shows data processingfor one forward channel burst. (The boundary between the upper and thelower physical layers is where the baseband signals are translated-intofrequency tones. The lower physical layer can then be regarded as thecommon element of the various modes and directions of transmission.) Thelarge shaded area shows the processing required for one traffic channelat the Base. The remainder of the diagram shows how various trafficchannels are combined. The details of each block in the diagram arediscussed throughout this section

[0282] The binary source delivers data to the Base transmitter at 64kbits/sec. This translates to 48 bits in one forward transmission burst.

[0283] The information bits are encrypted according to the triple dataencryption standard (DES) algorithm.

[0284] The encrypted bits are then randomized in the data randomizationblock. The bit to octal conversion block converts the randomized binarysequence into a sequence of 3-bit symbols. The symbol sequence isconverted into 16 symbol vectors. (In this description, the term vectorgenerally refers to a column vector. A vector is generally complexunless otherwise stated. Generally, column vectors are denoted by boldlower-case characters, while row vectors are denoted by the samecharacters with a transpose operation, denoted by a superscript T.Another widely used vector form used here is a conjugate transposevector referred to here as Hermetian.) One symbol from the LCC is addedto form a vector of 17 symbols.

[0285] The 17-symbol vector is trellis encoded. The trellis encodingstarts with the most significant symbol (first element of the vector)and is continued sequentially until the last element of the vector (theLCC symbol). This process employs convolutional encoding that convertsthe input symbol (an integer between 0 and 7) to another symbol (between0 and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is signal withinthe set of 16 QAM (or 16PSK) constellation signals. (The term signalwill generally refer to a signal constellation point.)

[0286] A link maintenance pilot signal (LMP) is added to form an18-signal vector, with the LMP as the first elements of the vector. Theresulting (18×1) vector d_(fwd) is pre-multiplied by a (18×18) forwardsmearing matrix C_(fwd-smear) to yield a (18×1) vector b.

[0287] Vector b is element-wise multiplied by the (18×1) gainpreemphasis vector g_(fwd)(p) to yield another (18×1) vector, c, where pdenotes the traffic channel index and is an integer in the range [0,M_(base)] where M_(base) is the maximum number of traffic channels thatcan simultaneously be carried over one traffic partition. Vector c ispost-multiplied by a (1×32) forward spatial and spectral spreadingvector g^(H) _(fwd)(p) to yield a (18×32) matrix R(p). The number 32results from multiplying the spectral spreading factor 4 and spatialspreading factor 8. The 18×32 matrices corresponding to all trafficchannels carried (on the same traffic partition) are then combined(added) to produce the resulting 18×32 matrix S_(fwd).

[0288] The matrix S_(fwd) is partitioned (by groups of four columns)into eight (18×4) submatrices (A₀ to A₇). (The indices 0 to 7,corresponds to the antenna elements over which these symbols willeventually be transmitted.) Each submatrix is mapped to tones within onetraffic partition (denoted by partition A in FIG. 31) according to themapping discussed in FIG. 22 and sent to the lower physical layer.

[0289] The lower physical layer places the baseband signals in discreteFourier transfer (DFT) frequency bins where the data is converted intothe time domain and sent to its corresponding antenna elements (0 to 7)for transmission over the air. The details of the lower physical layerare discussed below.

[0290] This process is repeated from the start for the next 48 bits ofbinary data to be transmitted in the next forward transmission burst.The various steps in the transformation of binary data are shown in FIG.32. To keep the diagram simple, the spreading and traffic channelcombiner functions are shown in one step.

[0291] Medium Capacity Mode

[0292] The block diagram for the upper physical layer of the Basetransmitter for the medium capacity mode is shown in FIG. 33. Theprimary difference between the transmission formats for high and mediumcapacity modes is the use of different trellis encoding schemes. Inmedium capacity mode, an 8QAM (or 8PSK) rate 2/3 trellis encoder(compared to a 16QAM or 16PSK rate 3/4 bits in one forward transmissionburst, two traffic partitions (A and B) are used.

[0293] The binary source delivers binary data to the Base transmitter at64 kbits/sec. For one forward channel burst, this translates to 48 bits.The information bits are encrypted according to the triple DESalgorithm. The encrypted bits are then randomized in the datarandomization block. The bit to-bit conversion block converts therandomized binary sequence into a sequence of 2-bit symbols. The symbolsequence is converted into 24 symbol vectors. Two symbols from the LCCare added, and eight ones are inserted at the end of the sequence toform a vector of 34 symbols. (The two symbols for the LCC carry onlythree bits of LCC information. The least significant bit, LSB, of thesecond LCC symbol is always set to one.)

[0294] The 34-symbol vector is trellis encoded. The trellis encodingstarts with the most significant symbol (first element of the vector)and is continued sequentially until the last element of the vector (thesecond LCC symbol). This process employs convolutional encoding thatconverts the input symbol (an integer between 0 and 3) to another symbol(between 0 and 7) and maps the encoded symbol to its corresponding 8 QAM(or 8PSK) signal constellation point. The output of the trellis encoderis therefore a vector of 34 elements where each element is a signalwithin a set of 8QAM (or 8PSK) constellation signals.

[0295] The 34-element vector is divided into two 17-element vectors. AnLMP is added to each of the vectors to form two 18-element vectors dfdand d′_(fwd), where the LMP is the first element of these vectors. Eachresulting vector is pre-multiplied by a (18×18) forward smearing matrixC_(fwd-smear) to yield another two (18×1) vector b and b′. Vectors b andb′ are then element-wise multiplied by two (18×1) gain preemphasisvectors g_(fwd)(p) and g′_(fwd)(p) to yield two (18×1) vectors c and c′,where p denotes the traffic channel index. Each vector ispost-multiplied by its corresponding (1×32) Forward Spatial and Spectralspreading Vector (g^(H) _(fwd)(p) or (g′^(H) _(fwd)(p)) to yield two(18×32 matrices R(p) and R′(p).

[0296] The various 18×32 matrices corresponding to all traffic channelscarried on traffic partition A are combined to produce the 18×32 matrixS_(fwd). Similarly, matrices from those traffic channels carried onTraffic Partition B are combined to produce the 18×32 matrix S′_(fwd).

[0297] Matrix S_(fwd) is partitioned (by groups of four columns) intoeight (18×4) submatrices (A₀ to A₇). Each submatrix is mapped into toneswithin partition A according to the mapping discussed in FIG. 22 and issent to the lower physical layer. Similarly, matrix S′_(fwd) ispartitioned into eight (18×4) submatrices (A′₀ to A′₇). Each submatrixis mapped into tones with partition B according to the mapping discussedin FIG. 22 and is sent to the lower physical layer.

[0298] The lower physical layer places the baseband signals in DFTfrequency bins where the data is converted into the time domain and sentto its corresponding antenna element (0 to 7) for transmission over theair.

[0299] This process is repeated from the start for the next 48 bits ofbinary data to be transmitted in the next forward channel transmissionburst. The various steps in the transformation of binary data are shownin FIG. 34. To keep the diagram simple, the spreading and trafficchannel combiner functions are shown in one step. The block diagram forthe upper physical layer of the Base transmitter for low capacity modeis shown in FIG. 35.

[0300] Low Capacity Mode

[0301] The primary difference between the transmission formats for highand low capacity modes is the use of different trellis encoding schemes.In low capacity mode, a rate 1/2 trellis encoder (compared to a rate 3/4encoder for high capacity mode) is employed. To transmit 48 bits in oneforward transmission burst, three Traffic Partitions (A, B, and C) areused.

[0302] The binary source delivers binary data to the Base transmitter at64 kbits/sec For one forward channel burst, this translates to 48 bits.The information bits are encrypted according to the Triple DESalgorithm. The encrypted bits are then randomized in the datarandomization block. The 48 bits are formed into a vector. Three symbolsfrom the LCC are then added to form a vector of 51 symbols. The51-symbol vector is trellis encoded. The trellis encoding starts withthe most significant symbol (first element of the vector) and iscontinued sequentially until the last element of the vector (the thirdLCC symbol). This process employs convolutional encoding that convertsthe binary input symbol (0 or 1) to another symbol (0, 1, 2, or 3) andmaps the encoded symbol to its corresponding QPSK signal constellationpoint. The output of the trellis encoder is therefore a vector of 51elements where each element is a signal within the set of QPSKconstellation signals.

[0303] The 51-element vector is divided into three 17-element vectors.An LMP is added to each of the vectors to form three 18-element vectorsd_(fwd), d′_(fwd), and d″_(fwd), where the LMP is the first element ofthese vectors. Each resulting vector is pre-multiplied by a (18×18)forward smearing matrix C_(fwd-smear) to yield another three (18×1)vectors b, b′, and b″. Vectors b, b′, and b″ are then element-wisemultiplied by their respective (18×1) gain preemphasis vectorsg_(fwd)(p), g′_(fwd)(p), and g″ to yield three (18×1) vectors c, c′, andc″, where p denotes the traffic channel index. Each vector ispost-multiplied by its corresponding (1×32) forward spatial and spectralspreading vector (g^(H) _(fwd)(p), g′^(H) _(fwd)(p), or g″_(fwd)(p)) toyield three (18×32) matrices R(p), R′(p), and R″(p).

[0304] The various 18×32 matrices corresponding to traffic channelscarried on traffic partition A are combined to produce the 18×32 matrixS_(fwd). Similarly, matrices from those traffic channels carried ontraffic partitions B and C are combined to produce two 18×32 matrices,S′_(fwd) and S″_(fwd), respectively. Matrix S_(fwd) is partitioned (bygroups of four columns) into eight (18×4) submatrices (A₀ to A₇). Eachsubmatrix is mapped into tones within partition A according to themapping discussed in FIG. 22 and is sent to the lower physical layer.Matrix S′_(fwd) is partitioned into eight (18×4) submatrices (A′₀ toA′₇). Each submatrix is mapped into tones within partition B and is sentto the lower physical layer. Similarly, matrix S″_(fwd) is partitionedinto eight (18×4) submatrices (A″₀ to A″₇). Each submatrix is mappedinto tones within partition C and is sent to the lower physical layer.The lower physical layer places the baseband signals in DFT frequencybins where the data is converted into the time domain and sent to itscorresponding antenna element (0 to 7) for transmission over the air.

[0305] This process is repeated from the start for the next 48 bits ofbinary data to be transmitted in the next forward channel transmissionburst. The various steps in the transformation of binary data are shownin FIG. 36. To keep the diagram simple, the spreading and trafficchannel combiner functions are shown in one step. Similarly, theencryption and randomization functions are also shown in one step.

[0306] Encryption/Decryption

[0307] The 64 kbps binary source delivers bits to the encryption module48 bits at a time The encryption function is a three-stage cascade ofthe DES algorithm as shown in FIG. 37.

[0308] Trellis Encoding/Decoding

[0309] The trellis encoding technique consists of convolutional encodingfollowed by a signal mapping. The three modes of the physical layer usedifferent trellis codes. For high capacity mode, there are two possiblesignal constellations: 16PSK and 16QAM.

[0310] The rate 3/4 convolutional encoder for the 16 PSK constellationis shown in FIG. 38. The convolutional encoder employs an 8-state(k=4)¹⁴ rate 1/2 mother encoder that encodes one bit out of a 3-bitinput symbol, and passes the remaining bits uncoded.

[0311] The rate 1/2 convolutional encoder for the 16PSK constellationmay be described by the generator polynomials (G₀=04, G₁=13), in octalrepresentation. Equivalently, the polynomial representation is:

G _(n) =D

G ₁ =D ³ +D ²+1

[0312] The rate 3/4 convolutional encoder for the 16QAM constellation isshown in FIG. 39. The convolutional encoder employs an 8-state (k=4)rate 1/2 mother encoder that encodes one bit out of a 3-bit inputsymbol, and passes the remaining bits uncoded.

[0313] The rate 1/2 convolutional encoder for the 16QAM constellationmay be described by the generator polynomials (G₀=17, G₁=13), in octalrepresentation. Equivalently, the polynomial representation is:

G ₀ =D ³ +D ² +D+1

G ₁ =D ³ +D ²1

[0314] The two highest-order bits of the input symbol (x₂, x₁) arepassed through uncoded to form the two highest-order bits of the outputsymbol (y₃, y₂). The lowest-order bits of the input symbol (x₀) entersthe rate 1/2 mother encoder (shown as the shaded box) to produce twolowest-order bits of the output symbol (y₁, y₀)

[0315] The next step in the trellis encoding process is to map theoutput symbol onto a signal in the 16QAM (or 16PSK) constellation Theparticular mappings for the 16QAM and 16PSK constellations are shown inFIG. 40.

[0316] The resulting trellis encoder output is one of 16 possiblecomplex numbers within the 16QAM (or 16PSK) constellation shown in FIG.40. The actual value of each constellation point (signal) is shown inFIG. 41. The points on the constellation have been chosen so that theaverage energy of the signal is 1.

[0317] In medium capacity mode, a rate 2/3 trellis code with either 8QAMor 8PSK signal mapping is employed. The convolutional encoder for the8PSK constellation is shown in FIG. 42. It employs a 32 state (k=6) rate1/2 mother encoder that encodes one bit out of a 2-bit input symbol, andpasses the remaining bit uncoded. The rate 1/2 convolutional encoder maybe described by the generator polynomials (G₀=10, G₁=45), in octalrepresentation. Equivalently, the polynomial representation is:

G ₀ =D ²

G ₁ =D ⁵ +D ³ +D ²+1

[0318] The convolutional encoder for the 8QAM constellation is shown inFIG. 43. It employs a 32-state (k=6) rate 1/2 mother encoder thatencodes one bit out of a 2-bit input symbol, and passes the remainingbit uncoded. The rate 1/2 convolutional encoder may be described by thegenerator polynomials (G₀=53, G₁=75), in octal representation.Equivalently, the polynomial representation is:

G ₀ =D ⁵ +D ⁴ D ²+1

G ₁ =D ⁵ +D ³ +D ²+1

[0319] The highest-order bit of the input symbol (x₁) is passed throughuncoded to form the highest-order bit of the output symbol (y₂). Thelowest-order bit of the input symbol (x₀) enters the rate 1/2 motherencoder to produce the two lowest-order bits of the output symbol (y₁,y₀).

[0320] The next step in the trellis encoding process is to map theoutput symbol onto a signal in the 8QAM (or 8PSK) constellation. Theparticular mappings for the 8QAM and 8PSK constellations are shown inFIG. 44. The resulting trellis encoder output is one of eight possiblecomplex numbers within the 8QAM (or 8PSK) constellation shown in FIG.44. The actual value of each constellation point (signal) is shown inFIG. 45. The points of the constellation have been chosen so that theaverage energy of the signal is 1.

[0321] In low capacity mode, the de facto standard rate 1/2 encoder,shown in FIG. 46, together with QPSK mapping is employed. The only bitof the input symbol (x₀) enters the rate 1/2 mother encoder to producethe two bits of the output symbol (y₁, y₀).

[0322] The next step in the trellis encoding process is to map theoutput symbol onto a signal in the QPSK constellation. The particularmapping for the QPSK constellation is shown in FIG. 46 referred to bynatural mapping. The resulting trellis encoder output is one of fourpossible complex numbers within the QPSK constellation shown in FIG. 47.The actual value of each constellation point (signal) is shown in FIG.48. The points on the constellation have been chosen so that the averageenergy of the signal is one.

[0323] Cluster Smearing/Desmearing

[0324] This section defines the smearing matrix C_(fwd-smear). The inputto the smearing block is (18×1) vector d_(fwd). The output of thesmearing operation (vector b) can then be described by the matrixmultiplication of d_(fwd) and the (18×18) smearing matrix C_(fwd-smear),That is

b=C _(fwd-smear) d _(fwd)

[0325] C_(fwd-smear) is the constant valued matrix shown below:$\left\lbrack \quad \begin{matrix}1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & {\beta\delta 0} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & {\beta\delta 1} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & {\beta\delta 2} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & {\beta\delta 3} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & {\beta\delta 4} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & {\beta\delta 5} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 6} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 7} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 8} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 9} & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 10} & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 11} & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 12} & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 13} & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 14} & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 15} & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 16}\end{matrix}\quad \right\rbrack\quad$

[0326] where,

a=(r _(LMP)/(1+r _(LMP)))^(½)

b=(1/(1+r _(LMP)))^(½)

[0327] and r_(LMP) is the ratio of pilot to data power that is aphysical layer provisionable parameter whose value is nominally set toone.

[0328] Gain Preemphasis

[0329] This section discusses the gain preemphasis matrix g_(fwd)(p)shown in FIG. 31. The input to the gain preemphasis block is the (18×1)vector b. The output of the gain preemphasis operation (Vector c) is theelement-wise multiplication of vector b and the gain preemphasis vectorg_(fwd)(p):

c=b·g _(fwd)(p)

[0330] where · represents element-wise vector multiplication. Theelements of g_(fwd)(p) are derived using information received at theBase. The derivation of these weights are implementation dependent.

[0331] Spectral and Spatial Spreading

[0332] This section defines the (1×32) forward spatial and spectralspreading vector g^(H) _(fwd)(p) shown in FIG. 31. The input to thespectral and spatial spreading block is the 18-element vector c. Theoutput of the spectral and spatial spreading operation, (18×32) matrixR(p), is the matrix multiplication of c and the (1×32) spectral andspatial spreading vector g^(H) _(fwd)(p):

R(p)=cg ^(H) _(fwd)(p)

[0333] where,

g ^(H) _(fwd)(p)=[g ₀ g ₁ g ₂ . . . g ₃₀ g ₃₁]

[0334] The elements of vector g^(H) _(fwd)(p) are transmit spreadingweights calculated throughout the transmission. The algorithm for thederivation of these weights is implementation dependent. However, toclarify the procedure, a specific algorithm for the derivation of theseweights is described below.

[0335] The Base derives its new weights based on the most recent datareceived on the reverse channel. The transmit weights are a scaledversion of the received weights using eight antenna inputs with fourreceive frequencies per antenna. The receive weight vector w^(H)_(rev)(p) has 32 elements (w₀-w₃₁) that are mapped to spatial andspectral components as shown in FIG. 49.

[0336] For the Base traffic establishment procedure, the transmitweights (g₀-g₃₁) are calculated according to the following equation:

g ^(H) _(fwd)(p)=a _(fwd)(n)h(k _(fwd) W ^(H) _(rev)(p))

[0337] where k_(fwd) is the Base transmission constant, a_(fwd)(n) isthe Base gain ramp-up factor for the nth packet, and h(.) is a functionthat limits the norm of its argument to 23 dBm

.h(v)=v

for ∥v∥ ²<23 dBm

h(v)=23 dBm (scale factor) (v/∥v∥ ²)

[0338] otherwise

[0339] For the Base steady-state procedure, receive weights areadaptively calculated using the following equation:

w _(rev)(p)=R ⁻¹ _(xx) r _(xy)

[0340] where

[0341] w_(rev)(p) is the (32×1) weight vector;

[0342] r_(xy) is an estimate of the (32×1) cross-correlation vector ofthe received (32×1) vector x and the despread data y, multiplied by anestimate of the channel equalization weights; and

[0343] R⁻¹ _(xx) is an estimate of (32×32) inverted auto-correlationmatrix of the received vector x. (R⁻¹ _(xx) may be computed using theRecursive Modified Gramm-Schmidt (RMGS) algorithm.)

[0344] r_(xy) is cross-correlated against the despread data after aresmearing step and a gain pre-emphasis reapplication step.

[0345] The receive weights (w₀-w₃₁) are mapped to spatial and spectralcomponents according to the mapping shown in FIG. 49. The transmitweights (g₀-g₃₁) are a scaled version of the receive weights. Thescaling is made according to the following equation:

g ^(H) _(fwd)(p)=k _(fwd) w ^(H) _(rev)(p)

[0346] where k_(fwd) is the Base steady-state transmission constant.

[0347] Correlation estimates are computed over eight reverse-channelbursts. The new despreading weights are applied to eight reverse channelburst with no delay. The spreading weights are applied to eight forwardchannel bursts after a 4-burst delay Correlation estimates are madeusing an exponentially averaged block summation. The exponential decayconstant is provisionable with a nominal value of 0.7.

[0348] An illustrative flowchart of an embodiment of the adaptivesolution of spectral and spatial weights is shown in FIG. 85.

[0349] Forward Control Channel Transmission Format

[0350] The block diagram for the physical layer of the Common LinkChannel (CLC) channel transmissions is shown in FIG. 50. A CLC messageis a 64-bit binary sequence. The bit to di-bit conversion block convertsthe binary sequence into a sequence of 2-bit symbols of length 32. Thevector formation block converts the symbol sequence into a (32×1)vector. Each element of the resulting vector is mapped into itscorresponding signal in the QPSK signal constellation to form another(32×1) vector, s. The mapping for the QPSK signal is shown in FIG. 51.

[0351] The resulting vector is passed through two parallel paths. In thefirst path, the vector s is sent directly for spectral and spatialspreading that involves post-multiplying it by the (1×32) spreadingvector g_(clc) ^(H):

g ^(H) _(clc) =[g 0 g 1 g 2 . . . g 30 g 31]

[0352] (g^(H) _(clc) is discussed further below.) The resulting (32×32)matrix is D_(clc). Matrix D_(clc) is then sent to the antennademultiplexer where it is partitioned (by groups of 4 columns)into eight(32×4) submatrices A₀ to A₇. The elements of these matrices willultimately be transmitted on antennas 0 to 7, respectively.

[0353] In the second path, the vector s is code-gated. The code-gatingoperation is described by the element-wise multiplication of the (32×1)vector s with a (32×1) code-gating vector Y_(clc). The resulting (32×1)vector is S′:

s′=s·i _(clc)

[0354] The vector i_(clc) is described below.

[0355] The resulting (32×1) vector s′ is sent for spectral and spatialspreading that involves post-multiplying it by the (1×32) spectral andspatial spreading vector g_(clc) ^(H). The resulting (32×32) matrix isD′_(clc) Matrix D′_(clc) is then sent to the antenna demultiplexer whereit is partitioned (by groups of 4 columns) into eight (32×4) submatricesA′₀ to A′₇. Each of these matrices (A₀ to A₇) and (A′₀ to A′₇) is thensent to a time demultiplexer where it is further partitioned (by groupsof 4 rows) into eight (4×4) submatrices. This yields 128 (4×4) matrices(D₀ to D₆₃) and (D′₀ to D′₆₃).

[0356] The transmission of one 64 bit CLC message requires 16 forwardchannel bursts or 4 TDD periods. In each of these bursts, eight (one foreach antenna) of the (4×4) matrices are mapped onto tones and sent tothe lower physical layer for transmission over the air. The interleavingand tone mapping functions are described herein.

[0357] The vector g_(clc) is defined as the Kronecker product of a (8×1)spatial spreading vector d and a (4×1) spectral spreading vector f:

g _(clc)=kron(d, f)

[0358] where d, is given by ${\begin{matrix}d_{0} \\d_{1} \\d_{2} \\d_{3} \\d_{4} \\d_{5} \\d_{6} \\d_{7}\end{matrix}}\quad$

[0359] and f is given by ${\begin{matrix}{f\quad 0} \\{f\quad 1} \\{f\quad 2} \\{f\quad 3}\end{matrix}}\quad$

[0360] The resulting vector g_(clc) is given by: ${\begin{matrix}{\delta \quad 0\quad f\quad 0} \\{{\delta 0}\quad f\quad 1} \\{{\delta 0}\quad f\quad 2} \\{{\delta 0}\quad f\quad 3} \\{{\delta 1}\quad f\quad 0} \\\vdots \\{{\delta 7}\quad f\quad 2}\end{matrix}}\quad$

[0361] The g_(clc) ^(H) is the conjugate transpose of g_(clc).

[0362] The spreading vector f is a column of the (4×4) Hadamard matrixH₄, that may be chosen randomly by the Base.

[0363] The spreading vector d is the kth column of the (8×72) CLCSpatial Spreading Weights Table. The column index k is provided by theMAC layer through the parameter CLC beam.

[0364] A (N×N) Hadamard matrix denoted by H_(N) is obtained by thefollowing recursion: H_(2n) equals ${\begin{matrix}{Hn} & {Hn} \\{Hn} & {Hn}\end{matrix}}\quad$

[0365] where H₀ is initialized at 1. For instance, the 4×4 Hadamardmatrix (H₄) is: ${\begin{matrix}1 & 1 & 1 & 1 \\1 & {- 1} & 1 & {- 1} \\1 & 1 & {- 1} & {- 1} \\1 & {- 1} & {- 1} & 1\end{matrix}}\quad$

[0366] The code-gating vector i_(clc) is:

i _(clc) =b _(clc) ·h _(clc)

[0367] where the vector h_(clc) is the 0th column of the (32×32)Hadamard matrix (H₃₂), that is the all ones vector and the ith elementof the (32×1) vector b_(clc) is given by:

b _(clc) =e ^(j2pik)offset^(/32)

[0368] The k_(offset) (an integer between 0 and 31) is the Base stationoffset code (BSOC) for the transmitting Base.

[0369] Interleaving

[0370] There are 16 burst in every CLC transmission (burst 0 to burst15). For each antenna, the interleaver outputs one of the 16 possible(4×4) matrices in each burst. FIG. 52 shows the order of thetransmission used by the interleaver.

[0371] Tone Mapping

[0372] There are 128 (4×4) matrices to be mapped onto tones fortransmission over the air. FIG. 53 shows the mapping of a (4×4) matrixat the output of the interleaver into tones. The absolute tone indicescan be obtained using FIG. 23.

[0373] The Broadcast Channel

[0374] The block diagram for the physical layer of BRC channeltransmissions is shown in FIG. 54. The block diagram is very similar tothat for the CLC shown in FIG. 50 However, for the sake of completeness,and to point out the small differences, the details of the BRCtransmission format are included in this section.

[0375] The primary differences between the CLC and the BRC transmissionson the forward channel are:

[0376] The Base uses all our BRC channels (in the four sub-band pairs)while for the CLC, channel selection is based on its operating sub-bandpair.

[0377] The Base forms 10 spatial beams (activated sequentially) to coverall the RUs in one hemisphere, that means that the time to broadcast aBRC message is ten times as long as transmission of a CLC message.

[0378] A BRC message is a 64-bit binary sequence. The bit to di-bitconversion block converts the binary sequence into a sequence of 2-bitsymbols of length 32. The vector formation block converts the symbolsequence into a (32×1) vector. Each element of the resulting vector ismapped into its corresponding signal in the QPSK signal constellation toform another (32×1) vector s. The mapping for the QPSK signal isidentical to that for the CLC shown in FIG. 51.

[0379] The resulting vector is passed through two parallel paths. In thefirst, path, the vector s is sent directly for spectral and spatialspreading that involves post-multiplying it by the (1×32) spectral andspatial spreading vector g_(brc) ^(H.).

g ^(H) _(brc) =[g 0 g 1 g 2 . . . g 30 g 31]

[0380] The g_(brc) ^(H) is discussed below.

[0381] The resulting (32×32) matrix is D_(brc). Matrix D_(brc) is thensent to the antenna demultiplexer where it is partitioned (by groups offour columns) into eight (32×4) submatrices A₀ to A₇. The elements ofthese matrices will ultimately be transmitted on antennas 0 to 7,respectively.

[0382] In the second path, the vector s is code-gated. Code-gating isdescribed by the element-wise multiplication of the (32×1) vector s witha (32×1) code-gating vector i_(brc). The resulting (32×1) vector is s′:

s′=s·Y _(brc)

[0383] The vector Y_(brc) is described below.

[0384] The resulting (32×1) vector s′ is sent for spectral and spatialspreading, that involves post-multiplying it by the (1×32) spreadingvector g_(brc) ^(H). The resulting (32×32) matrix is D′_(brc). MatrixD′_(brc) is then sent to the antenna demuitiplexer where it ispartitioned (by groups of four columns) into eight (32×4) submatricesA′₀ to A′₇. Each of these matrices (A₀ to A₇) and (A′₀ to A′₇) is thensent to a time demultiplexer where they are further partitioned (bygroups of four rows) into eight (4×4) submatrices. This yields 128 (4×4)matrices (D₀ to D₆₃) and (D′₀ to D′₆₃).

[0385] For one spatial beam, the transmission of one 64-bit BRC messagerequires 16 forward channel bursts or four TDD periods. In each of thesebursts, eight (one for each antenna) of the (4×4) matrices are mappedonto tones and sent to the lower physical layer for transmission overthe air. The interleaving and tone mapping functions are describedherein.

[0386] This process is repeated 10 times to provide 10 spatial beamswith different directions so that all the RUs in a spatial cell candetect the broadcast message. The details of the beam sweeping aredescribed below. The duration of a BRC transmission is therefore 160bursts or 40 TDD periods.

[0387] The vector g_(brc) is defined as the Kronecker product of a (8×1)spatial spreading vector d and a (4×1) spectral spreading vector f:

g _(brc)=kron(d, f)

[0388] where d, is given by ${\begin{matrix}d_{0} \\d_{1} \\d_{2} \\d_{3} \\d_{4} \\d_{5} \\d_{6} \\d_{7}\end{matrix}}\quad$

[0389] and f is given by ${\begin{matrix}{f\quad 0} \\{f\quad 1} \\{f\quad 2} \\{f\quad 3}\end{matrix}}\quad$

[0390] The resulting vector g_(brc) is given by ${\begin{matrix}{\delta \quad 0\quad f\quad 0} \\{{\delta 0}\quad f\quad 1} \\{{\delta 0}\quad f\quad 2} \\{{\delta 0}\quad f\quad 3} \\{{\delta 1}\quad f\quad 0} \\\vdots \\{{\delta 7}\quad f\quad 2} \\{{\delta 7}\quad f\quad 3}\end{matrix}}\quad$

[0391] The g_(brc) ^(H) is the conjugate transpose of g_(brc). Thespreading vector f is a column of the (4×4) Hadamard matrix H₄, that maybe chosen randomly by the Base. The spreading vector d is a column ofthe BRC Spatial Spreading Weights Table that will be described in thenext release of this document. The Base transmits simultaneously on allthe sub-band pairs; for each sub-band pair, 10 different spatial beamsare formed and activated sequentially to cover al the RUs in the spatialcell.

[0392] The code-gating vector i_(brc) is:

i _(brc) =b _(brc) h _(brc)

[0393] where the vector h_(brc) is the 0th column of the (32×32)Hadamard matrix (H₃₂), that is the all ones vector and the ith elementof the (32×1) vector b_(clc) is given by:

b _(brc) =e ^(j2pik)offset³²

[0394] The k_(offset) (an integer between 0 and 31) is the BSOC for thetransmitting Base.

[0395] For every spatial beam, there are 16 bursts in every BRCtransmission (burst 0 to burst 15). For each antenna, the interleaveroutputs one of the 16 possible (4×4) matrices in each burst. Theinterleaving rule is identical to the CLC interleaving rule shown inFIG. 52. There are a total of spatial beams. This process is thereforerepeated sequentially ten times, once for every spatial beam.

[0396] For every spatial beam, there are 128 (4×4) matrices to be mappedonto tones for transmission over the air. FIG. 55 shows the mapping of a(4×4) matrix at the output of the interleaver into tones. The absolutetone indices can be obtained using FIG. 23.

[0397] Broadcast channel signals are spatially beamformed andtransmitted sequentially via ten predetermined team patterns persub-band pair. This results in four broadcast channel signals (one persub-band pair) that are simultaneously swept through each spatial cell.This is shown in FIG. 56.

[0398] Each BRC message requires 40 TDD intervals or 120 ms to transmit.New BRC message may only be started on even frame boundaries. Each ofthe four sub-band pairs transmits the same BRC message at the same timeand the BRC beam sweeps are synchronous within a spatial cell and forall Bases within a this embodiment system. BRC beams are swept in aclockwise pattern.

[0399] Reverse Channel Format

[0400] As for the forward channel transmissions, there are threedifferent possible implementations of the physical layer. We refer tothese modes as:

[0401] High capacity mode (short range): 3 bits/symbol

[0402] Medium capacity mode (medium range): 2 bits/symbol

[0403] Low capacity mode (long range): 1 bit/symbol.

[0404] High Capacity Mode

[0405] The block diagram for the upper physical layer of the Remote Unittransmitter for high capacity mode is shown in FIG. 57.

[0406] The binary source delivers binary data to the Remote Unittransmitter at 64 kbits sec. For one reverse channel burst, thistranslates to 48 bits. The information bits are encrypted according tothe triple DES algorithm. The encrypted bits are then randomized in thedata randomization block.

[0407] The bit to octal conversion block converts the randomized binarysequence into a sequence of 3-bit symbols. The symbol sequence isconverted into 16-symbol vectors One symbol from the LCC is added toform a vector of 17 symbols.

[0408] The 17-symbol vector is trellis encoded. The trellis encodingstarts with the most significant symbol (first element of the vector)and is continued sequentially until the last element of the vector (theLCC symbol). This process employs convolutional encoding that convertsthe input symbol (an integer between 0 and 7) to another symbol (between0 and 15) and maps the encoded symbol to its corresponding 16QAM (or16PSK) signal constellation point. The output of the trellis encoder istherefore a vector of 17 elements where each element is a signal withinthe set of 16QAM (or 16PSK) constellation signals.

[0409] A LMP signal is added to form an 18-signal vector, with the LMPas the first element of this vector. The resulting vector d_(rev) ispre-multiplied by a (18×18) reverse smearing matrix C_(rev-smear) toyield a (18×1) vector b. Vector b is post-multiplied by a (1×4) reversespreading vector g^(H) _(rev) to yield a (18×4) matrix S_(rev). Elementsof matrix S_(rev) are mapped to tones within traffic partition Aaccording to the mapping discussed in FIG. 22 and are sent to the lowerphysical layer. The lower physical layer places the baseband signals intheir corresponding DFT frequency bins where the data is converted intothe time domain and sent for transmission over the air.

[0410] This process is repeated from the start for the next 48 bits ofbinary data to be transmitted in the next reverse channel transmissionburst. The various steps in the transformation of binary data are shownin FIG. 58.

[0411] Medium Capacity Mode

[0412] The block diagram for the upper physical layer of the Remote Unittransmitter for medium capacity mode is shown in FIG. 59.

[0413] The binary source delivers binary data to the Remote Unittransmitter at 64 kbits/sec. For one reverse channel burst, thistranslates to 48 bits. The information bits are encrypted according tothe triple DES algorithm. The encrypted bits are then randomized in thedata randomization block. The bit to di-bit conversion block convertsthe randomized binary sequence into a sequence of 2-bit symbols. Thesymbol sequence is converted into 24 symbol vectors. Two symbols fromthe LCC are added, and eight ones are inserted at the end of thesequence to form a vector of 34 symbols.

[0414] The 34-symbol vector is trellis encoded. The trellis encodingstarts with the most significant symbol (first element of the vector)and is continued sequentially until the last element of the vector (thesecond LCC symbol). This process employs convolutional encoding thatconverts the input symbol (an integer between 0 and 3) to another symbol(between 0 and 7) and maps the encoded symbol to its corresponding 8 QAM(or 8PSK) signal constellation point. The output of the trellis encoderis therefore a vector of 34 elements where each element is a signalwithin the set of 8QAM (or 8PSK) constellation signals.

[0415] The 34-element vector is divided into two 17-element vectors. AnLMP is added to each of the vectors to form two 18-signal vectorsd_(rev) and d′_(rev), with the LMP as the first element of thesevectors. Each resulting vector is pre-multiplied by a (18×18) reversesmearing matrix C_(rev-smear) to yield another two (18×1) vectors b andb′. Each vector is post-multiplied by its corresponding (1×4) reversespreading vector (g^(H) _(rev) or g′^(H) _(rev)) to yield two (18×4)matrices S_(rev) and S′_(rev). Elements of matrix S_(rev) are mapped totones within traffic partition A according to the mapping discussed inFIG. 22 and are sent to the lower physical layer. Similarly, elements ofmatrix S′_(rev) are mapped to tones within traffic partition B and aresent to the lower physical layer. The lower physical layer places thebaseband signals in DFT frequency bins where the data is converted intothe time domain and sent for transmission over the air.

[0416] This process is repeated from the start for the next 48 bits ofbinary data to be transmitted in the next reverse channel transmissionburst. The various steps in the transformation of binary data are shownin FIG. 60.

[0417] Low Capacity Mode

[0418] The block diagram for the upper physical layer of the Remote Unittransmitter for low capacity mode is shown in FIG. 61. The binary sourcedelivers binary data to the Remote Unit transmitter at 64 kbits/sec. Forone reverse channel burst, this translates to 48 bits. The informationbits are encrypted according to the triple DES algorithm. The encryptedbits are then randomized in the data randomization block. The 48 bitsare formed into a vector, and three bits from the LCC are added to forma vector of 51 bits.

[0419] The 51-bit vector is trellis encoded. The trellis encoding startswith the most significant bit (first element of the vector) and iscontinued sequentially until the last element of the vector (the thirdLCC bit). This process employs convolutional encoding that converts thebinary input symbol (0 or 1) to another symbol (0, 1, 2, or 3) and mapsthe encoded symbol to its corresponding QPSK signal constellation point.The output of the trellis encoder is therefore a vector of 51 elementswhere each element is a signal within a set of QPSK constellationsignals. The 51-element vector is divided into three 17-element vectors.An LMP is added to each of the vectors to form three (18×1) vectorsd_(rev), d′_(rev), and d″_(rev), with the LMP as the first element ofthese vectors. Each resulting vector is pre-multiplied by a (18×18)reverse smearing matrix C_(rev-smear) to yield another three (18×1)vectors b, b′, and b″. Each vector is post-multiplied by itscorresponding (1×4) reverse spreading vector (g^(H) _(rev), g′^(H)_(rev), or g″^(H) _(rev)) to yield three (18×4 matrices S_(rev),S′_(rev), and S″_(rev). Elements of matrix S_(rev) are mapped to toneswithin traffic partition A according to the mapping discussed in FIG. 22and are sent to the lower physical layer. Similarly, elements ofmatrices S′_(rev), and S″_(rev) are mapped to tones within trafficpartition B and C, respectively, and are sent to the lower physical. Thelower physical layer places the baseband signals in their correspondingDFT frequency bins where the data is converted into the time domain andsent for transmission over the air.

[0420] This process is repeated from the start for the next 48 bits ofbinary data to be transmitted in the next reverse channel transmissionburst. The various steps in the transformation of binary data are shownin FIG. 62.

[0421] The encryption function is identical to that for the forwardchannel described herein.

[0422] The trellis encoding schemes for all three capacity modes areidentical to those for the forward channel described herein.

[0423] This section specifies the smearing matrix C_(rev-smear). Theinput to the smearing block is the (18×1) vector D_(rev). The output ofthe smearing operation (vector b) can then be described by the matrixmultiplication of d_(rev) and the (18×18) smearing matrix C_(rev-smear).That is

b=C _(rev=smear) d _(rev)

[0424] C_(fwd-smear) is the constant valued matrix shown below:$\left\lbrack \quad \begin{matrix}1 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & {\beta\delta 0} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & {\beta\delta 1} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & {\beta\delta 2} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & {\beta\delta 3} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & {\beta\delta 4} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & {\beta\delta 5} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 6} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 7} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 8} & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 9} & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 10} & 0 & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 11} & 0 & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 12} & 0 & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 13} & 0 & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 14} & 0 & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 15} & 0 \\\alpha & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & 0 & {\beta\delta 16}\end{matrix}\quad \right\rbrack\quad$

[0425] where,

a=(r _(LMP)/(1+r _(LMP)))^(½)

b=(1/(1+r _(LMP)))^(½)

[0426] r_(LMP) is the ratio of pilot to data power that is a physicallayer provisionable parameter whose value is nominally set to one.

[0427] The d, s are elements of the cluster scrambling vector d_(smear)that is unique to the Remote Unit. d_(smear) is a 17-element vector thatis used to ensure that the smeared data from one user received in aparticular traffic partition at the Base is uncorrelated with otherusers within the same traffic partition in the local spatial cell andadjacent spatial cells. d_(smear) is given by: ${\begin{matrix}d_{0} \\d_{1} \\\vdots \\d_{15} \\d_{16}\end{matrix}}\quad$

[0428] or ${\begin{matrix}{e\quad j\quad {\Phi (0)}} \\{e\quad j\quad {\Phi (1)}} \\\vdots \\{e\quad j\quad {\Phi (15)}} \\{e\quad j\quad {\Phi (16)}}\end{matrix}}\quad$

[0429] The ith element of d_(smear) has the form e^(jf) _(smear) ^((i))where _(fsmear)(i) is a real number between 0 and 2 p generated by apseudo-random number generator creating unique sequences for each RemoteUnit. The details of the pseudo-random number generator areimplementation dependent and need not be known at the Base.

[0430] Spectral Spreading

[0431] This section defines the (1×4) reverse spectral spreading vectorg^(H) _(rev) shown in FIG. 57. The input to the spectral spreading blockis the (18×1) vector b. The output of the spectral and spatial spreadingoperation, (18×4) matrix S_(rev), is the matrix multiplication of b andthe (1×4) Spectral spreading vector g^(H) _(rev).

S _(rev) =bg ^(H) _(rev)

[0432] where,

g ^(H) _(rev) =[g 0 g 1 g 2 . . . g 30 g 31]

[0433] The elements of vector g^(H) _(rev) are transmit spreadingweights calculated throughout the transmission. The algorithm for thederivation of these weights is implementation dependent. However, toclarify the procedure a specific algorithm for the derivation of theseweights is described below.

[0434] The Remote Unit derives its new transmit weights based on themost recent data received on the forward channel. The transmit weightsare a scaled version of the received weights using four receivefrequencies for a single antenna.

[0435] The receive weight vector w^(H) _(fwd) has four elements (w₀-w₃)that are mapped to spectral components as shown in FIG. 63.

[0436] For the Remote Unit traffic establishment procedure, the transmitweights (g₀-g₃) are calculated according to the following equation:

g ^(H) _(rev)(p)=a _(rev)(n)p _(rev) w ^(H) _(fwd)

[0437] where a_(fwd)(n) is the Base gain ramp-up factor for the nthpacket and where p_(rev) is the Remote Unit power management factordefined by the equation below:

p _(rev)=1_(p) k _(fwd)+(1−1_(p))k _(rev)(p_(loss)(n, p)/∥(w _(fwd)(p))∥

[0438] where,

[0439] 1_(p) is the exponential decay or “forget factor” nominally setto 0.97

[0440] p_(loss) is the reciprocal of the Base-Remote Unit channel gainmeasured using the Remote Unit Synchronization Pilot (RSP) tones

[0441] k_(rev) is the target Base receive power (nominally −103 dBm)

[0442] n is the burst index

[0443] p is the link index.

[0444] For the Remote Unit traffic establishment procedure, the receiveweights are adaptively calculated using the following equation:

w _(fwd) =R ⁻¹ _(xx) r _(xd)

[0445] where

[0446] w_(fwd) is the (4×1) receive weight vector

[0447] r_(xd) is an estimate of the (4×1) cross-correlation vector ofthe received (4×1) vector x and the LMP (or the desired data) d

[0448] R⁻¹ _(xx) is an estimate of the (4×4) inverted auto-correlationmatrix of the received vector x

[0449] For the Remote Unit steady-state procedure, receive weights areadaptively calculated using the following equation:

w _(fwd) =R ⁻¹ _(xx) r _(xy)

[0450] where

[0451] w_(fwd) is the (4×1) weight vector

[0452] r_(xy) is an estimate of the (4×1) cross-correlation vector ofthe received (4×1) vector x and the despread data y

[0453] R⁻¹ _(xx) is an estimate of the (4×4) inverted auto-correlationmatrix of the received vector x.

[0454] The receive weights (w₀-w₃) are mapped to spectral componentsaccording to the mapping shown in FIG. 63. The transmit weights (g₀-g₃)are a scaled version of the receive weights. The scaling is madeaccording to the following equation:

g ^(H) _(rev)(p)=p _(rev) w ^(H) _(fwd)

[0455] where p_(rev) is the Remote Unit power management factor definedearlier.

[0456] Correlation estimates are computed over four forward-channelburst. The new despreading weights are applied to four forward channelbursts with no delay. The spreading weights are applied to eight reversechannel bursts after an 8-burst delay. Correlation estimates are madeusing an exponentially average block summation. The exponential decayconstant is provisional with a nominal value of 0.7.

[0457] Reverse Control Channel Transmission Format

[0458] The block diagram for the physical layer of the solicited andunsolicited Common Access Channel (CAC) channel transmissions is shownin FIG. 64.

[0459] A CAC message is a 56-bit binary sequence composed of a trainingsequence, information bits, and CRC parity bits. The vector formationblock converts the binary sequence into a (56×1) vector. Each element ofthe resulting vector is mapped into its corresponding signal in the BPSKsignal constellation to form another (56×1) vector s. The mapping forthe BPSK signal is shown in FIG. 65.

[0460] The resulting vector is passed through two parallel paths. In thefirst path, the vector s is sent directly for spectral spreading thatinvolves post-multiplying it by the (1×2) spectral spreading vectorg_(cac) ^(H):

g ^(H) _(cac)=[1 1]

[0461] The resulting (56×2) matrix is D_(cac) given by:${\quad \begin{matrix}{s(0)} & {s(0)} \\{s(1)} & {s(1)} \\\cdots & \cdots \\{s(54)} & {s(54)} \\{s(55)} & {s(55)}\end{matrix}\quad }\quad$

[0462] where s(k) is the kth element of vector s. Matrix D_(cac) is thensent to the demultiplexer where it is partitioned (by group of eightrows) into seven (8×2) submatrices D₀ to D₆.

[0463] In the second path, the vector s is code-gated. The code-gatingoperation is described by the element-wise multiplication of the (56×1)vector s with a (56×1) code-gating vector Y_(cac). The resulting (56×1)vector is s′:

s′=s·i _(cac)

[0464] The vector i_(cac) is described below.

[0465] The resulting (56×1) vector s′ is sent for spectral spreadingthat involves post-multiplying it by the (1×2) spectral spreading vectorg_(cac) ^(H). The resulting (56×2) matrix is D′_(cac).${\quad \begin{matrix}{s^{\prime}(0)} & {s^{\prime}(0)} \\{s^{\prime}(1)} & {s^{\prime}(1)} \\\cdots & \cdots \\{s^{\prime}\left( 5 \right.} & {s^{\prime}\left( 5 \right.} \\{s^{\prime}\left( 5 \right.} & {s^{\prime}\left( 5 \right.}\end{matrix}\quad }\quad$

[0466] where s′(k) is the kth element of vector s′. Matrix D′_(cac) isthen sent to the demuitiplexer where it is partitioned (by groups ofeight rows) into seven (8×2) submatrices D′₀ to D′₆.

[0467] The transmission of one 56 bit CAC message requires 14 reversechannel bursts. In each of these burst, one of the 14 (8×2) matrices ismapped onto tones and sent to the lower physical layer for transmissionover the air. The interleaving and tone mapping functions are describedbelow.

[0468] The code-gating vector i_(cac) is:

i _(cac) =b _(cac) ·h _(cac)

[0469] and

b _(cac)(I)=e ^(j2pik)offset^(/56)

[0470] where b_(cac)(i) is the ith element of the (56×1) vector b_(cac).The k_(offset) is the BSOC for the receiving Base, that ranges between 0and 31. Every Remote Unit is assigned a pair of code keys: the solicitedCAC code key and the unsolicited CAC code key. The code keys are integernumbers between 0 and 63.

[0471] The 56 elements of the vector h_(cac) are the first 56 elementsof the kth column of the (64×64) Hadamard matrix (H₆₄), where k is thevalue of the solicited or the unsolicited code key for the transmittingRemote Unit depending on the type of CAC transmission. For instance, if,for a given Remote Unit, the solicited code key is the number 13, andthe unsolicited code key is the number 15:

[0472] In SCAC transmissions, elements of the vector h_(cac) are thefirst 56 elements of the 13th column of the (64×64) Hadamard matrix.

[0473] In UCAC transmissions, elements of the vector h_(cac) are thefirst 56 elements of the 15th column of the (64×64) Hadamard matrix.

[0474] There are 14 burst in every CAC transmission (burst 0 to burst13). The interleaver outputs one of the 14 possible (8×2) matrices (D₀to D₆) or (D′₀ to D′₆), in each burst. FIG. 66 shows the order of thetransmission used by the interleaver. There are two CACs in eachsub-band pair. The Remote Unit will use one of these channels dependingon the CAC ID parameter received from its MAC layer. If the CAC ID is 0,the CAC_(i, 0) is selected; if the CAC ID is 1, the CAC _(i, 1) isselected. FIG. 67 shows the mapping of the (8×2) matrix at the output ofthe interleaver into tones.

[0475] Lower Physical Layer Format

[0476] The transmitter for the lower physical layer of this embodimentis functionally described by the block diagram in FIG. 68. The lowerphysical layer functionality is identical in forward and reversechannels.

[0477] In the forward channel, for traffic channel transmissions, theprocess shown in FIG. 68 is performed in parallel eight times for eightdifferent antenna elements. Furthermore, the Base may combine dataintended for various users into the same DFT bin to reduce processingrequirements. It is possible to further reduce the processing bysimultaneously transmitting traffic and control information (at the Baseor the Remote Unit) as they are carried on non-overlapping frequencytones. These techniques, however, are implementation dependent and donot change the functional characteristics of the DFT operation. As shownin FIG. 68, complex baseband signals enter the tone mapping block, wherethey are assigned into tones according to a unique mapping to either atraffic or a control channel.

[0478] The tone-mapped complex signals are complex signals aredemultiplexed into lower sub-band and upper sub-band tones, and areplaced into their corresponding DFT bins. The remaining DFT bins arefilled with zeros and the inverse DFT operation is performed, therebytransforming the data into the time domain. The discrete time-domainsamples are then converted into an analog signal, converted to theappropriate RF frequency, and transmitted over the antenna.

[0479] As there are four sub-band pairs, there are four pairs of DFTblocks, where each DFT block spans one MHz of usable bandwidth. Thespacing between the adjacent bins in one DFT block is 3.125 kHz. EachDFT block has 512 bins of which only 320 bins are used. Tone mappinginto corresponding DFT bins in each DFT block are shown in FIG. 69. FIG.70 depicts tone mapping pictorially. As shown, the frequency span of oneDFT block is 1.6 MHz where only 1 MHz is used for data transmission. Therelationship between tones and the actual frequency for each bin isexplained herein.

[0480] The inverse DFT operation is carried out to convert the basebandsignals into time domain. The mathematical representation of theoperation is:

x(n)=S X(k)^(ej2pnk/512)

[0481] where X(k) is the complex baseband signals in the frequencydomain (the contents of the kth bin of a DFT block), and x(n) is the nthreal-valued component of the time-domain sample. The inverse DFToperation may be carried out using Inverse Fast Fourier Transform (IFFT)techniques.

[0482] The baseband transmit signals obtained after the IDFT operationmust be real. The real-valued time-domain sample outputs are thenconverted to the proper RF frequency and the appropriate analog waveformfor transmission.

[0483] Airlink Physical Layer Power Output Characteristics

[0484] The power output characteristics of Base transmissions on theforward channel are different from that of the Remote Unit transmissionson the reverse channel.

[0485] The forward channel transmission from a Base to a given RemoteUnit is maintained at a fixed power level during the duration of aconnection. The power level is determined by the Base radio managemententity (RME) prior to the start of the connection using a powermanagement algorithm.

[0486] A forward RF channel transmission is initiated by a 180 msramp-up period (240 forward channel bursts) during the trafficestablishment period. The ramp-up starts after a connection isestablished between the Base and a given Remote Unit. The datatransmitted during this period are known link maintenance pilots. Themaximum (steady state) power is reached after 240 channel bursts (180msec) and maintained throughout the connection.

[0487] The following equation shows the forward channel ramp-up schedulerelative to the steady state power,

a _(fwd)(n)=(1−e ^(−5(8[n/8]))/(1−e ⁻⁵))²

for n<240

a _(fwd)(n)=1

[0488] otherwise

[0489] where n is the forward channel burst number relative to the startof the transmission.

[0490] The reverse channel transmissions from a Remote Unit to its Baseis adaptively varied to ensure that the received power from all RUs attheir Base is maintained at a relatively constant level. The Remote Unitpower management algorithm is implementation dependent. One example ofthe algorithm is discussed in the Section on the Reverse Channel Format.

[0491] A reverse RF channel transmission is initiated by a 180 msramp-up period (240 reverse channel bursts) during the trafficestablishment period. The ramp-up starts after a connection isestablished between the Remote Unit and its Base. The data transmittedduring this period are known LMPs. The maximum (steady state) power isreached after 240 reverse channel bursts (180 msec).

[0492] The following equation shows the reverse channel ramp-up schedulerelative to the steady state power,

a _(rev) ⁻(n)=(1−e ^(−5(8[n/8]))/(1−e ⁻⁵))²

for n<240

a _(rev)(n)=1

[0493] otherwise

[0494] where n is the reverse channel burst number relative to the startof the transmission.

[0495] A “Proof-of-Concept” Embodiment

[0496] The signal processing procedure described generally above can beimplemented in a “proof-of-concept” embodiment by circuitry within thehigh-bandwidth base station 110 and the radio access stations 187, 192.In addition, the dynamic bandwidth allocation method of the presentinvention is implemented in a “proof-of-concept” embodiment within thecircuitry of the communications network 100 depicted below.

[0497]FIG. 71 is a schematic block diagram showing the main structuralelements of one implementation of the high bandwidth efficiency,bandwidth-on-demand communications network 100. Specifically, thecommunications network 100 is shown to include a plurality of full-rate,high-bandwidth, radio access stations 192 as well as a low-rate,high-bandwidth radio access station 187. Typically, a full-rate,high-bandwidth radio access station 192 is able to provide forcommunication between a base station 110 and a large number ofsubscribers 130, while the low-rate, high-bandwidth radio access station187 is able to provide for communication with the base station 110 onlyone or a few subscribers 130 at a time.

[0498] The subscribers 130 communicate with the full-rate or low-rate,high-bandwidth radio access stations 192, 187 via a cable or othercommunication link. The high-bandwidth radio access stations 192, 187,in turn, communicate bidirectionally with the base station 110 viawireless communications channels to form an air-link. The structure andoperation of the base station 110 as well as the structure and operationof the full and low-rate, high-bandwidth radio access stations 192, 187,will be discussed in greater detail below with reference to FIG. 72 and73.

[0499] The base station 110 together with the full and low-rate,high-bandwidth radio access stations 192, 187 together comprise asubsystem 150. The subsystem 150 communicates bidirectionally with atelecommunications network 160 via a land line 170 that may, forexample, comprise a copper cable or a fiber-optic connection.Alternatively, the link 170 may comprise a microwave link. Thetelecommunications network 160 may include for example, the publicswitched telephone network, a mobile telephone switching office (MTSO),a private data network, a modem bank or a private branch exchange, as iswell known in the art.

[0500]FIG. 74 is a simplified schematic block diagram that shows themain functional and structural elements of the bandwidth-on-demandcommunications network 100 in greater detail. The communications network100 is shown in FIG. 74 to connect a plurality of the subscriber units(e.g., the computer 131, the telephone 132, a plurality of telephones140 in communication with a public switching network or a plurality ofcomputer terminals 145 within a local area network) to public or privatedata or telephone networks 150-156. Public data network 150, the privatedata network 152, the private telephone network 154, and the publictelephone network 156 communicate with an asynchronoustelecommunications multiplexer (ATM) 162 over lines 163, 164, 165, and166, respectively, via a plurality of network interfaces, designatedgenerally by a block 160. The asynchronous telecommunicationsmultiplexer 162 acts as a multiplexing switch and connects with thehigh-bandwidth base station 192 via the communications link 170, thatadvantageously comprises a fiberoptic link, a copper wire, or amicrowave transmission link. The high-bandwidth base station 110provides radio frequency output signals to the receiving stations viathe antenna 120.

[0501] A bandwidth demand controller 175 communicates with thehigh-bandwidth base station 110, asynchronous telecommunications managerswitch 162, and the network interfaces 160 via lines 176, 177 and 178,respectively. The bandwidth demand controller 175 also communicates withan intelligent service node 180 via a line 179. The intelligent servicenode 180 communicates with the ATM switch 162 via a line 182. Theabove-described elements of the bandwidth on demand communicationsnetwork 100 comprise the telecommunications network side 183 of thebandwidth-on-demand communication system 100. The telecommunicationsnetwork side 183 communicates with the low rate high-bandwidth radioaccess station 187 via an antenna 185 or the full rate high band widthradio access station 192 via an antenna 190. The radio access station187 connects to a plurality of the subscriber units including thetelephone 132 and the computer 131. The radio access station 192 isconfigured to communicate with multiple subscribers 140 via a publicswitching network 195 that connects to the radio access station 192 viaa communications link 194. The full rate radio access station 192further connects to the computer terminals 145 via a local network 197that connects to the radio access station 192 via communication link196. Each of the subscriber units 131, 132, 140 and 145, together withthe elements 185-197 of the communications system 100, comprise asubscriber network side 199 of the bandwidth-on-demand communicationssystem 100.

[0502] The high-bandwidth base station 110 in association with thebandwidth demand controller 175, and the high-bandwidth radio accessstations 187, 192 that communicate with the high-bandwidth base station110 via an air link, are the heart of the bandwidth on demandcommunication system 100. Although only a single high-bandwidth basestation 100 is depicted in FIG. 74, it will be understood that aplurality of high-bandwidth base stations are advantageously includedwithin the high-bandwidth communication system 100. Each of thehigh-bandwidth base stations 110 is capable of supporting from one tohundreds of simultaneous bi-directional users. Each user may request inadvance, or optionally, during the course of communications, an amountof bandwidth from 8 kilobits to 1.544 megabits per second. Furthermore,each high-bandwidth base station 110 may have one or multipletransmitting and receiving antennas 120. The high-bandwidth basestations 110 are high-bandwidth radio transceivers, that, with theirassociated antennas 120, may be located on towers, on top of buildings,inside buildings, or in other convenient locations.

[0503] A bandwidth controller 175 is associated with the high-bandwidthbase stations 110. The bandwidth demand controller 175 providesintelligence to monitor information transmitted to the base stations 110from the radio access stations 187, 192. Specifically, the informationtransmitted from the high-bandwidth radio access stations 187, 192 areconverted to intelligence within the bandwidth demand controller 175 inorder to instruct the base stations 110 how much bandwidth to provide agiven radio access station 187, 192. Although shown in FIG. 74 as aseparate element from the high-bandwidth base station 110, the bandwidthdemand controller 175 may be integral to a base station 110, may beattached locally to a base station 110, or may be remote and connectedto a base station 110 via the communication link 176. The bandwidthdemand controller 175 further acts as a central bandwidth controllerthat insures that the bandwidth appropriated at each communication linkthroughout the communications network 183 is consistent with thebandwidth assigned to a particular channel on the high-bandwidth basestation 110. Thus, the bandwidth demand controller 175 controlsbandwidth allocated within the asynchronous telecommunicationsmultiplexer switch 162, and the network interfaces 160. Additionally,the bandwidth demand controller 175 communicates bandwidth informationto the intelligent service node 180 that is used to manage the deliveryof the user data to the appropriate network 150-156. The intelligentservice node 180 can control the ATM switch 162 to manage bandwidthchanges and the network interfaces 160.

[0504] The high-bandwidth radio access stations 187, 192, as shown inFIG. 74, are exemplary of a plurality of high-bandwidth radio accessstations that are included within the bandwidth on demand communicationsystem 100. One or more of the high-bandwidth radio access stations 187,192 are capable of communicating with one or more high-bandwidth basestations 110 utilizing the air interface. In addition, each of the radioaccess stations 187, 192 is capable of supporting one or more interfacessuch as the connections between the telephone 132 and the standardcomputer 131, as well as the telephone network interface (PBX) 195 andthe computer network comprising the terminals 145 and the LAN 197. Thehigh-bandwidth radio access stations 187, 192 have the capability tointerpret bandwidth needs of the devices connected to the radio accessstations 187, 192, and communicate these bandwidth needs via the airinterface and the base station 110, to the bandwidth demand controller175. Advantageously, the bandwidth demand controller 175 can furthercommunicate these bandwidth demands to the ATM switch 162 or theintelligent service node 180, and the network interfaces 160.

[0505] In operation, one of the connected subscriber units (e.g., thecomputer 131, the telephone 132, the PBX 195, or the LAN 197) requestsbandwidths via a connection to one of the high-bandwidth radio accessstations 187, 192. The radio access station 187, 192 transmits a requestfor access and bandwidth to the high-bandwidth base station 110 via theantenna 185, 190, the air interface, and the antenna 120. The requestfor access is made via a communications control channel available to allsubscribers within the area of use. If two subscribers simultaneouslyrequest connection, then a random accessing protocol is employed todetermine which unit is first granted control of the communicationscontrol channel.

[0506] The high-bandwidth base station 110 communicates all bandwidthrequests to the bandwidth demand controller 175. The bandwidth demandcontroller performs an allocation of the requested bandwidth andadvantageously arranges system resources within the telecommunicationsnetwork side 183 (including the intelligent service node 180, the ATMswitch 162, and the network interfaces 160). Once the bandwidth demandcontroller 175 determines the amount of bandwidth available forallocation, and compares this with the requested bandwidths, thebandwidth demand controller 175 either immediately allocates therequested bandwidth, or begins a negotiation process using the availableamount of bandwidth. This bandwidth negotiation occurs between thebandwidth demand controller and the radio access station 187, 192through the base station 110 and the air interface.

[0507] Thus, the radio access station 187, 192 either receives anacknowledgment that the bandwidth requested is available, andsubsequently begins transmitting data, or the radio access station 187,192 receives an offer of less bandwidth from the bandwidth demandcontroller 175. If an offer of less bandwidth is transmitted to thehigh-bandwidth radio access station 187, 192, the radio access station187, 192 determines whether the connected device or network caneffectively operate with the offered bandwidth. If the connected deviceor network can effectively operate with the offered bandwidth, the radioaccess station 187, 192 begins transmitting data at the offeredbandwidth. However, if the radio access station 187, 192 determines thatthe offered bandwidth is not adequate for operation of the connecteddevice or network, the radio access station 187, 192 notifies theconnected device or network that access is not available, and furthernotifies the bandwidth demand controller 175 (via the base station 110and the air interface) that the offered bandwidth will not be used bythe radio access station 187, 192.

[0508] If a suitable bandwidth is available, the bandwidth controllerallocates this bandwidth to establish a communications channel with therequesting subscriber. Thus, for example, the telephone subscriber unit132 may indicate that a data rate of 8 Kb per second is required (thatcorresponds to a particular bandwidth) while the computer subscriberunit 131 may indicate that a total transmission rate of 128 Kb persecond (corresponding to another given bandwidth) in order to establisheffective communications with the high-bandwidth base station 110. Ifthe communications network 100 is unable to provide the requested amountof bandwidth, a negotiations process commences wherein thehigh-bandwidth base station 110 transmits an alternative bandwidth, thatis less than the requested bandwidth, to the requesting subscriber unitvia the radio access station 187. The requested subscriber unit thenindicates to the high-bandwidth base station 110 whether or not theallocated bandwidth is suitable for the communications needs of thesubscriber unit.

[0509] As will be described in greater detail below, the bandwidthdemand controller 175 allocates bandwidth by assigning one or morefrequency tone set and one or more spreading code to the subscriber unitin accordance with a pre-defined bandwidth allocation procedure. Eachtone set and spreading code increases bandwidth by an additional factor.In one advantageous embodiment, bandwidth can be allocated in amounts assmall as 8 Kbits/sec to as large as 1.544 Mbits/sec to define thecommunications channel.

[0510] Once a communications channel is established for the requestingsubscriber 130, data representing either human voice communications orcomputer-to-computer communications in digital form, is transmittedbetween the high-bandwidth base station 110 and the high-bandwidth radioaccess station 187, 192. As will be described in greater detail below,the digitally encoded signal contains forward error correction togetherwith signal spreading and other modulation techniques.

[0511] All data received by the high-bandwidth base station 110 from allcommunicating radio access stations 187, 192 are multiplexed into anasynchronous telecommunications multiplexed data stream and transmitted,via the communication link 170, to the ATM switch 162. At the ATM switch162, the data stream is switched (i.e., demultiplexed) with the optionalassistance of the intelligent service node 180 to the appropriatenetwork interfaces 160, and from there onto the appropriate network150-156.

[0512] As will be discussed in further detail below, the bandwidthdemand controller 175 also controls bandwidth allocation for the networkinterfaces 160 and the ATM switch 162. In this manner, the bandwidthallocated throughout an entire communications link (i.e., from asubscriber to a data or telephone network) can be flexibly assignedaccording to the needs of each subscriber unit. Furthermore, thepreferred embodiment assures that the bandwidth through the airinterface and the bandwidth through the land line connections areappropriately matched.

[0513] When a device or network connected to a high-bandwidth radioaccess station 187, 192 no longer requires bandwidth, the radio accessstation 187, 192 ceases transmission to the base station 110, andnotifies the bandwidth demand controller that the bandwidth is nowreleased for reallocation.

[0514] The “Proof-of-Concept Embodiment”—Remote Terminal Hardware

[0515]FIG. 72 is a functional block diagram that shows the mainfunctional elements of the full-rate, high-bandwidth radio accessstation 192. It should be understood, for purposes of the presentdescription, that the fill-rate, high-bandwidth radio access station 192described herein is substantially identical in structure and operationto the low-rate, high-bandwidth radio access station 187, with theexception that the low-rate, high-bandwidth radio access station 187provides communication access for only a single subscriber 130. As shownin FIG. 72, the full-rate, high-bandwidth radio access station 192comprises a transmit receive switch 300 that connects bidirectionallywith the antenna 120. The structure and operation of the antenna 120will be described in greater detail below with reference to FIGS. 6 and7. The transmit and receive switch 300 connects to a down converter 305when in the receive mode and an up converter 307 while in a transmitmode. The transmit/receive switch 300 further receives synchronizationand packet timing data from a synchronization circuit 312.

[0516] The down converter 305 receives the radio signals from theantenna 120 via the switch 300. In addition, the down converter 305receives a local oscillator reference as well as an analog-to-digitalconverter clock from the synchronization circuit 312. The down converter305 communicates with a demodulator 310 that, in turn, provides afeedback of automatic gain control level to the down converter 305. Thedemodulator 310 communicates bidirectionally with the synchronizationcircuit 312 and also provides an output to a code-nulling circuit 315.The code-nulling circuit 315 provides a frequency error signal to thesynchronization circuit 312, and also communicates with amultidimensional trellis decoder 320. The multidimensional trellisdecoder 320 connects to a digital data interface 325. The digital datainterface 325 communicates bidirectionally with a remote control circuit330. The remote control circuit 330 receives inputs from the demodulatorcircuit 310, the code-nulling circuit 315, and the multidimensionaltrellis decoder 320. The control circuit 330 further transmits statussignals and receives command signals from the base station 110 (see FIG.74). Finally, the remote control circuit 330 outputs axis parameters toa multidimensional trellis encoder 335, that also communicates with thedigital interface 325. The multidimensional trellis encoder 335communicates with a SCMA coding circuit 340. The SCMA coding circuit 340further receives an input from the code-nulling circuit 315. The SCMAcoding circuit 340 outputs signals to a modulator circuit 345 that alsoreceives an input from the synchronization circuit 312. Finally, themodulation circuit 345 together with the synchronization circuit 312provide inputs to the up converter 307. The up converter 307 outputs thedata signal to the transmit receive switch 300 while the transmitreceive switch is in the transmit mode. This signal is output over anerror interface to the multiple subscribers 130 via the antenna 120.

[0517] In operation, once it is the proper time to receive a datapacket, the transmit/receive switch 300 switches the antenna 190 intothe down converter 305. The down converter 305 takes the signal at thetransmission frequency (see, e.g., about 2 gigaHertz), and translatesthis to the proper frequency for digitization. The DMT-SS demodulatorthen performs a fast Fourier transform (FFT) ard presents the individualfrequency bins to the code-nulling network 315. As discussed brieflyabove, the code-nulling network 315 applies code-nulling weights to thedespreading codes in order to cancel interference due to transmissionshaving non-orthogonal spreading codes. The code-nulling network 315 alsodespreads the demodulated signal provided by the DMT-SS demodulator 310and produces output demodulated symbols.

[0518] The demodulated symbols are provided as an input to themulti-dimensional trellis decoder 320 in order to decode the symbols inaccordance with pragmatic Viterbi decoding methods. Receive bits areprovided at the output of the multidimensional trellis decoder 320. Thereceive bits pass through a digital data interface 325 that, in oneembodiment, serves as a data interface for a T1 link.

[0519] On the transmit side, data to be transmitted enters the digitaldata interface 325 via the T1 link and enters the multidimensionaltrellis encoder 335 for trellis encoding. It will be understood, ofcourse, that other kinds of error encoding and symbol encoding such asReed-Solomon error coding, and QAM or BPSK symbol encoding are performedwithin the encoder 335. The encoded symbols enter the spreading circuit340 wherein the spreading code together with the appropriate codeweights are applied to the input symbols. The spread symbols are DMT-SSmodulated as represented within the block 345 and the resulting signalis translated to the high frequency band via the up-converter 307. Thetransmit/receive switch 300 is then switched to connect the up converter307 to the antenna 190 so that the modulated and encoded data signal istransmitted via the antenna 190.

[0520] Immediately after one of the radio access terminals 187, 192 hasbeen installed and is just coming on-line for the first time, the radioaccess station 187, 192 does not have information regarding the locationof the assigned base station 110. Furthermore, the remote access station187, 192 does not have information concerning the interference resultingfrom other transmitters and reflectors within the environment of theremote station 187, 192. Thus, each remote, upon initialization, must“learn” the location of the base station as well as the location ofdifferent interferes and reflectors within the immediate environment ofthe remote. Because the remote installer points the remote antenna arrayin the direction of the nearest base station 110, the strongest signalreceived by the remote is generally from around the 0° direction. Theremote subsequently fine tunes, or adaptively adjusts the beam formingso as to obtain the maximum SINR for the signal received from thenearest base station 110.

[0521] When the radio access station 192 transmits to the base 110, thebase station 110 expects to receive each of the signals transmitted fromthe remotes at the same power level. Thus, a gain control level isreported to the remote control 330 within the radio access station 192from the DMT-SS demodulator 310. This automatic gain control level(AGCL) is also transmitted from the DMT-SS modulator 310 to theup-converter 307 so that the gain of the power amplifier (not shownwithin the up-converter 307) can be adjusted. In this manner, the basestations 110 can assure that the signal transmitted from the remoteaccess terminals 187, 192 arrive at the base station 110 at the properlevel.

[0522] The radio access stations 187, 192 also have to performsynchronization. That is, although the remote access terminals 187, 192are preprogrammed to operate within a TDD system, the specificinformation concerning the distinction between the transmit and receivepackages as well as the exact timing of the packet transfer still mustbe determined by the radio access stations 187, 192 when a radio accessstation first comes on line. Subsequently, the remote terminals 187, 192must acquire frequency synchronization for the DMT-SS signals so thatthe remotes are operating at the same frequency and phase as the basestation 110. For this reason, the DMT-SS demodulator 310 generates apacket reference that is utilized by the synchronization circuitry 312to establish the basic transmit/receive timing (i.e., the packet timingfor the T/R switch 300). In addition, the packet timing is provided as areceive gate to the demodulator 310 and as a transmit gate to themodulator 345 so that the remote access station 187, 192 transmits andreceives at the appropriate intervals.

[0523] Within the code-nulling network 315, measurements are taken onthe waveform to determine the frequency error. The measured frequencyerror is provided to the synchronization circuitry 312 so that the radioaccess station 187, 192 can come into frequency and phase lock with thebase station 110. This synchronization information is transmitted fromthe synchronization circuitry 312 to the up-converter 307 and thedown-converter 305 as a local oscillator reference and also as adigital-to-analog converter clock (or conversely, an analog-to-digitalconverter clock).

[0524] The code-nulling network 315 also estimates the characteristicsof the multitask channel (i.e., the frequency response of the multipathchannel). The channel estimates are provided to the spreading circuitry340 so that the preemphasis function can be performed to adaptivelyequalize the multipath channel. Furthermore, the code-nulling network315 provides an estimate of the received power and the SINR to theremote control circuitry 330. In addition, an estimate of the bit errorrate (BER) is provided from the multidimensional trellis decoder 320 tothe remote control circuitry 330. These parameters are used by theremote control circuitry 330 to control the flow of data via the digitaldata interface 325 with the subscriber (e.g., a PBX or a LAN).Furthermore, status signals based upon these input parameters to theremote control circuitry 330 are also transferred to the subscribers.The status signals indicate to the subscribers whether or not the radioaccess terminal is operating properly.

[0525] When the radio access terminals 187, 192 first dials onto thenetwork 100 (i.e., the remote is trying to establish connection with thebase) the base provides the remote 187, 192 with a set of accessparameters that includes, for example, the appropriate starting codes touse, which tone sets to receive and transmit on, etc., so that acommunication channel is set up between the base station 110 and theremote station 187, 192.

[0526]FIGS. 21A and 21B depicts the digital architecture within theremote access terminals 187, 192. The remote digital architectureincludes an interface card 2100 that communicates bidirectionally with alayer processing accelerator (LPA) card 2110 as well as a transmittingLPA card 2120.

[0527] The interface card 2100 (shown in greater detail in FIG. 76below) includes an ETHERNET interface card, a global positioning system(GPS) interface and other control interfaces. The ETHERNET interfacecommunicates bidirectionally with a monitoring computer such as an AppleMacIntosh, while the GPS interface derives timing data forsynchronization purposes from the base station transmission, while thecontrol interface outputs printer control bits for controlling thetuner. The interface card 2100 further includes three digital signalprocessing chips that, advantageously comprise PMS320C40 digital signalprocessing chips (“C40s”) available from Texas Instruments. In addition,a Viterbi decoder as well as a T1 and an integrated services digitalnetwork (ISDN) interface are included on the interface card 2100 toprovide an interface between the T1 communication link as well as theISDN communication link with the subscribers.

[0528] As shown in FIG. 75A, the interface card 2100 further includes anadditional PMS320C40 digital signal processing chip as well as anadditional Viterbi, T1, ISDN interface that are crossed out. This is toindicate that these chips, although physically present on the interfacecard 2100, are not used within the remote digital subsystem although thesame interface card is typically used in the base station 110. This isdone because it is less expensive to manufacture a single interface cardfor both the base station 110 and the remotes 187, 192 rather thanproviding a specific card for the remotes and bases.

[0529] Finally, the interface card 2100 includes a G-link receiver thatreceives sampled data from the receiver digital-to-analog converter anda G-link transmitter that transmits sample data to the transmitteranalog-to-digital converter.

[0530] The sample data received from the digital-to-analog converterpasses through the G-link receiver within the interface card 2100. TheG-link receiver provides the received waveform data to a receiving LPAcard 2110 (FIG. 75B). The LPA card 2110 will be described in greaterdetail below with reference to FIGS. 77A-77D. Briefly, the receiving LPAcard 2110 includes a pair of SHARP LH9124 (“9124s”) digital signalprocessing chips 2112, 2114, as well as a pair of Texas InstrumentsTMS320C40 digital signal processing chips 2116, 2118.

[0531] The receiving LPA card 2110 demodulates the received data andprovides the demodulated data to one of the TMS320C40 DSP chips withinthe interface card 2100. After further digital signal processing, thedata is decoded and then transmitted to the subscriber via the T1interface. Of course, it will be understood that if the radio accessterminal comprises one of the low-rate radio access terminals 187, thena suitable communications link other than a T1 link will connect to theinterface card 2100.

[0532] When data is to be transmitted, information supplied by the T1interface, or other communication link, enters the interface card 2100,as shown in FIG. 75A, and passes through a series of digital signalprocessing chips within the interface card 2100. The transmit dataoutput from the interface card 2100 enters a transmit LPA card 2120,that has a substantially similar architecture to the received LPA card2110. The transmit LPA card 2120 converts the transmit data intotransmit waveform data suitable to be sent to the transmitteranalog-to-digital converter via the G-link transmitter within theinterface card 2100.

[0533]FIG. 76 is a software block diagram that indicates the generalprocessing steps performed by each of the digital signal processingchips within the digital signal processing architecture of the radioaccess terminals 187, 192. Specifically, control signals are generatedby the TMS320C40 digital signal processing chips 2102, 2106, while thesymbol modulation (e.g., including trellis coded, Reed-Solomon, and QAM,BPSK, or M-ARY modulation) is performed by the digital signal processingchips 2104, 2106.

[0534] Within the receiving LPA, the 9124 digital signal processor 2112in conjunction with the C40 digital signal processor 2116 perform theoperations relating to the fast Fourier transform. The 9124 digitalsignal processor 2114 in conjunction with the C40 digital signalprocessor chip 2118 perform the processing steps relating to thecode-nulling and adaptive equalization aspects of the present invention.In like manner, within the transmitting LPA 2120, the C40 digital signalprocessing chip 2124, together with the 9124 digital signal processingchip 2128, perform the digital signal processing steps relating to theinverse fast Fourier transform (IFFT), while DSP chips 2122 and 2126perform the signal spreading operations used to provide modulation inaccordance with the present invention.

[0535] FIGS. 78A-78C are more detailed block diagrams showing thedigital architecture used to support the main digital signal processingC40 chips on the interface card 2100 of the remote terminals 187, 192.Several interface support circuits are employed to precondition datareceived by the digital signal processing chip 2102. In particular, areceive/transmit control interface circuit, an ETHERNET interfacecircuit, an erasable programmable logic device (EPLD) synchronizationcircuit, and a universal asynchronous receiver/transmitter (UART) serveas an interface between the DSP chip 2102 and circuitry external to theinterface card 2100. In addition, a programmable read-only memory(PROM)/random access memory (RAM), an electrically erasable PROM, areceived signal strength indicator (RSSI) input circuit and a pluralityof light-emitting diode (LED) switch drivers all communicate with theDSP chip 2102 via a common bus.

[0536] The C40 DSP chip 2104 is also supported by interface circuitry.Specifically, an integrated services digital network (ISDN) interfaceand a T1 interface provide a connection to ISDN and T1 equipment, whilea supporting Viterbi encoder/decoder, as well as a PROM/RAM, providedigital signal processing support for the DSP chip 2104 via a commonbidirectional bus.

[0537] In addition to receiving signals from the DSP chip 2104 andorderwire FFT data, the C40 DSP chip 2106 communicates bidirectionallywith a PROM/RAM and a codec via a bidirectional common bus. The codec incommunication with the DSP 2106 communicates bidirectionally with anorderwire headset.

[0538] The G-link receiver provides a clock synchronization signal tothe G-link transmitter, as well as to an EPLD. In addition, thereceiving G-link transmits RSSI data to the RSSI input in communicationwith the C40 DSP chip 2102. The EPLD that receives the synchronizationsignal from the receiving G-link circuit provides a receive address, aframe sync signal, and a transmit address as control outputs.

[0539] In operation, each of the DSP chips 2102, 2104, 2106 uses thelocal PROM/RAM for storage and retrieval of data and for use as alook-up table. The C40 DSP chip 2102 receives the RSSI input data toimplement automatic game control (AGC). That is, an indication of thesignal intensity is provided via the RSSI input to the DSP chip 2102 sothat the remote terminal 187, 192 can automatically adjust the receivegain so that the signal is received at the appropriate level. TheETHERNET interface allows the remote terminal 187, 192 to transmit dataout to a local computer or operator. The receive/transmit controlinterface circuit sends control bits to the radio frequency electronicsof the remote terminal 187, 192 in order to control the RF electronics.The EPLD synchronization circuit receives an envelope detector outputfrom the RF circuit within the receiver of the remote terminal 187, 192in order to achieve TDD synchronization. The UART circuit provides forthe input of a universal global positioning system (GPS) time clock foruse by the remote terminals 187, 192. Finally, the electrically erasablePROM allows the radio access terminals 187, 192 to store informationfrom test to test as a kind of statistical record.

[0540] The operations of the other support circuitry depicted in FIGS.78A-78C are well known to those of ordinary skill in the art and neednot be described in detail for a complete understanding of the presentinvention.

[0541] FIGS. 77A-77D are a more detailed block diagram of the LPA cards2110, 2120 of the remote terminals 187, 192 that shows the supportcircuitry used to support the operation of the SHARP LH9124 DSP chips,as well as the TMS320C4 DSP chips from Texas Instruments. It should beunderstood that although FIGS. 77A-77D depict only the receiving LPAcard 2110 that the architecture of the LPA card 2110 is substantiallysimilar to that of the transmitting LPA card 2120 so that essentiallythe same description applies to both LPA cards. Input data in quadratureform (e.g., 24 In-phase bits and 24 Quadrature bits) are provided as aninput to a double buffer 2402 via a 48-bit input bus. A first portion ofthe double buffer 2402 is controlled via input address and control bits,while a second portion of the double buffer 2402 is controlled via anaddress generator 2404. The address generator 2404 communicates with theTMS 320C40 DSP chip 2116 via a bus 2406.

[0542] The double buffer 2402 communicates with the SHARP LH9124 digitalsignal processing chip 2112 via a bidirectional bus and also suppliesdata as an input to a first in/first out (FIFO) buffer 2408. In onepreferred embodiment, the FIFO comprises a 5K×48-bit buffer. The FIFO2408 communicates with the DSP chip 2112, as well as with a doublebuffer 2410. Like the double buffer 2402, the double buffer 2410advantageously comprises a pair of 32K×48-bit RAMs. Furthermore, thedouble buffer 2410 is under the control of the address generator 2412that communicates with the buffer 2406. The double buffer 2410communicates bidirectionally with the DSP chip 2116 via the bus 2406.

[0543] The SHARP DSP chip 2112 further receives input from a sine/cosinelook-up table 2414. The sine/cosine look-up table 2414 receives inputfrom a rectangular-to-polar converter 2416 that in one embodimentcomprises a signal processing chip sold under Model Number PDSP16330 andavailable from GEC Plessey. Finally, the DSP chip 2112 receivessequencing data from a sequencer 2418, that also communicates with thebus 2406. The output of the digital signal processor chip 2112 isprovided as an input to a double buffer 2420, that is substantiallysimilar in structure to the double buffers 2402 and 2410. A firstportion of the double buffer 2420 is under the control of an addressgenerator 2422 that receives signals from the DSP chip 2116 via the bus2406.

[0544] The second half of the LPA card 2110 is substantially similar inarchitecture to the first half described above. Specifically, input datain quadrature form (e.g., 24 In-phase bits and 24 Quadrature bits) areprovided as an input to a second half of the double buffer 2420 from thefirst half of the buffer 2420. The second half of the double buffer 2420is controlled via an address generator 2424. The address generator 2404communicates with the TMS 321C40 DSP chip 2118 via a bus 2426.

[0545] The double buffer 2420 communicates with the SHARP LH9124 digitalsignal processing chip 2114 via a bidirectional bus and also suppliesdata as an input to a first in/first out (FIFO) buffer 2428. In onepreferred embodiment, the FIFO 2428 comprises a 5K×48-bit buffer. TheFIFO 2428 communicates with the DSP chip 2112, as well as with a doublebuffer 2430. Like the double buffer 2420, the double buffer 2430advantageously comprises a pair of 32K×48-bit RAMs. Furthermore, thedouble buffer 2430 is under the control of the address generator 2432that communicates with the buffer 2426. The double buffer 2430communicates bidirectionally with the DSP chip 2118 via the bus 2426.

[0546] The SHARP DSP chip 2114 further receives input from a sine/cosinelook-up table 2434. The sine/cosine look-up table 2434 receives inputfrom a rectangular-to-polar converter 2436 that in one embodimentcomprises a signal processing chip sold under Model Number PDSP16330available from GEC Plessey. Finally, the DSP chip 2114 receivessequencing data from a sequencer 2438, that also communicates with thebus 2426. The output of the digital signal processor chip 2114 isprovided as an input to a buffer 2440, that advantageously comprises a32K×48 RAM. The buffer 2440 is under the control of an address generator2442 that receives signals from the DSP chip 2118 via the bus 2426.

[0547] The C40 DSP chips 2116 and 2118, respectively, receive GPS timingvia UART circuits 2450, 2452. Furthermore, each of the DSP chips 2116,2118 communicates with respective RAM chips 2454, 2456, thatadvantageously comprise 128K×32 random access memories.

[0548] The DSP chips 2116, 2118 further communicate with EPROMs 2460,2470, respectively, and RAMs 2462, 2472, respectively, via local buses2464, 2474, respectively. In one advantageous embodiment the EPROMs2460, 2470 comprise a 512K×8 memory, while the RAMs 2462, 2472 comprisea 128K×32 RAM. A pair of internal communication ports provide forcommunication between the DSP circuits 2116, 2118, while two pair ofinput/output external communication ports connect to each of the DSPchips 2116, 2118.

[0549] In operation, the DSP chips 2116, 2118 employ the respectivememories 2460, 2462, 2470, 2472 to perform processing associated withthe fast Fourier transform and code spreading or code-nulling processingoperations. Meanwhile, the double buffer 2402 collects input datasymbols in quadrature. The double buffer 2402 is provided so that whiledata is being collected from one packet, data from the previous packetcan be processed.

[0550] As can be seen from FIGS. 77A-77D, two substantially identicalprocessing engines are provided separated by the double buffer 2420. Inone advantageous embodiment each of the 9124 DSPs 2112, 2114 operate ata 40-mHz sample rate and include six multipliers so that data can bestreamed through in substantially real time.

[0551] The “Proof-of-Concept Embodiment”—Base Station Hardware

[0552]FIG. 73 is a functional block diagram showing the main functionalelements of the base station 110 shown in FIG. 74. As shown in FIG. 73,the base station 110 includes a transmit/receive switch 400 thatcommunicates bidirectionally with a plurality of the antennas 120. Whilein the receive mode, the switch 400 communicates with a down converter405, and in the transmit mode, the transmit receive switch 400communicates with an up converter 407. The down converter 405 alsoreceives inputs from a frequency reference circuit 409 and providesoutputs to a demodulator 410. The demodulator 410 feeds back anautomatic gain control level to the down converter 405 and also receivesinputs from a packet timing generator 412. The packet timing generator412 receives analog-to-digital converter clock inputs from the frequencyreference circuit 409.

[0553] The demodulator 410 provides inputs to a beam forming andcode-nulling circuit 415. The beam forming and code-nulling circuit 415communicates with a multidimensional trellis decoder 420 that, in turn,communicates bidirectionally with a network/data interface circuit 425.

[0554] The network/data interface circuit 425 provides outputs to andreceives inputs from the telecommunications network 160 (see FIG. 74).Furthermore, the network/data interface circuit 425 provides an outputsignal to the packet timing generator 412 and also communicatesbidirectionally with a base control circuit 430. The base controlcircuit 430 receives inputs from the demodulator 410, the beamforming/code-nulling circuit 415, and the multidimensional trellisdecoder 420. The base control circuit 430 also communicatesbidirectionally with an operator station (not shown) within thetelecommunications network 160.

[0555] The network/data interface circuit 425 communicates with amultidimensional trellis encoder 435. The multidimensional trellisencoder 435 provides an output to a retroactive beam forming network andSCMA circuit 440. The network 440 also receives inputs from the beamforming/code-nulling circuit 415 as well as the base control circuit430. The retroactive beam forming and SCMA network 440 provides anoutput to a modulator 445 that also receives inputs from the packettiming generator 412. Finally, the modulator 445 together with thefrequency reference circuit 409 provide inputs to the up converter 407,that in turn provides an output to the transmit/receive switch 400 whilein the transmit mode. Signals provided by the up converter aretransmitted to the various high-bandwidth radio access stations 192, 187by means of the antennas 120.

[0556] The operation of a base station is substantially similar to theoperation of the radio access station 187, 192. Specifically, thetransmit/receive switch 400 switches the antenna array 120 into thedown-converter 405. The down-converter 405 takes the signal at thetransmission frequency (see, e.g., about 2 gigaHertz), and translatesthis to the proper frequency for digitization. The multi-sensor DMT-SSdemodulator 410 then performs a fast Fourier transform (FFT) andpresents the individual frequency bins to the beam forming/code-nullingnetwork 415. As discussed briefly above, the code-nulling network 415applies code-nulling and beam forming weights to the despreading codesin order to cancel interference due to transmissions havingnon-orthogonal spreading codes. The code-nulling network 415 alsodespreads the demodulated signal provided by the multi-sensor DMT-SSdemodulator 410 and produces output demodulated symbols.

[0557] The demodulated symbols are provided as an input to themulti-dimensional trellis decoder 420 in order to decode the symbols inaccordance with pragmatic Viterbi decoding methods. Receive bits areprovided at the output of the multidimensional trellis decoder 420. Thereceive bits pass through a digital data interface 425 that, in oneembodiment, serves as a data interface for a T3/SONET interface link.

[0558] On the transmit side, data to be transmitted enters the digitaldata interface 425 via the T3/SONET link and enters the multidimensionaltrellis encoder 435 for trellis encoding. It will be understood, ofcourse, that other kinds of error encoding and symbol encoding such asReed-Solomon error coding, and QAM or BPSK symbol encoding are performedwithin the encoder 435. The encoded symbols enter thebeam-forming/code-spreading circuit 440 wherein the spreading codetogether with the appropriate beam forming and null-steering codeweights are applied to the input symbols. The spread symbols are DMT-SSmodulated as represented within the block 445 and the resulting signalis translated to the high frequency band via the up-converter 407. Thetransmit/receive switch 400 is then switched to connect the up converter407 to the antenna array 120 so that the modulated and encoded datasignal is transmitted via the antenna 120.

[0559] For synchronization of the base stations 110 all of the bases 110are locked onto GPS time. In this manner, no matter how big thecommunications network 100 becomes, all of the base stations 110 alwayshave the proper TDD synchronization. Thus, the base stations 110 alwaysstart transmitting at the same time and receiving at the same time. Atthe packet timing generator 409, the frequency reference is GPS derivedand this is used to control the T/R switch 400. This is particularlyadvantageous because the timing does not have to be derived from thewaveforms transmitted by several remotes. Since the remote terminals187, 192 derive their synchronization timing from the base stations 110,the remotes will be synchronized to GPS time.

[0560] The packet timing generator 412 receives a clock signal from thetiming generator 409 so that the packet timing generator 412 can supplytransmit and receive gating signals to the modulator 445 and thedemodulator 410, respectively.

[0561] In an alternative embodiment, it would be possible to establish auniversal timing mechanism for all of the base stations 110 and theremote terminals 187, 192 provided from the network via the networkinterface 425. In such an embodiment, an specially defined ATMadaptation layer could be used to provide a clock to the interface 425.Management information and connection power control information couldalso be supplied over the T3 or SONET link. Such information could beprovided to the base controller 430 that will send the proper signalsout to the remotes 187, 192 through the wireless signaling network forconnection set up and carry out other management functions.

[0562] It should further be noted that the down-converter 405 andup-converter 407 contain separate RF electronics that include slightimperfections so that they may not be perfectly matched. For thisreason, a transmit/receive compensation is performed using additionalcompensation weights. The purpose of this compensation is to compensatefor the differences in phase and amplitude introduced into the signalsby the transmit and receive RF electronics. By applying the compensationweights, the same beam pattern is produced on the transmit side as onthe receive side.

[0563] FIGS. 79A-79D are a schematic block diagram that depicts theoverall digital signal processing architecture layout within the basestations 110. The base station 110 is laid out into a radio frequencychassis portion 2500 and a digital chassis portion 2510. The multipleelement antenna array 120, that for ease of illustration is depicted inFIGS. 79A-79D as comprising four antennas, connects to correspondingtransmit/receive modules 2512. Each transmit/receive module 2512includes the transmit/receive switch 400, as well as a receiver, atransmitter, and an amplifier. It should be noted that in accordancewith one advantageous aspect of the present invention, each antennaelement is provided with an individual amplifier. By using thisdistributed amplifier configuration instead of one large amplifier topower the entire antenna array, power is saved. In addition, in theevent of amplifier failure, only one of multiple antenna elements failsrather than the entire antenna array. Thus, the present inventionprovides for graceful degradation of signal quality in the event of anamplifier failure.

[0564] An analog-to-digital converter/digital-to-analog converter pair2515 provides for analog-to-digital and digital-to-analog conversion ofthe received and transmitted signals. The digitized received signalsenter the digital chassis 2510, while the digital transmit signals areprovided as an output of the digital chassis 2510.

[0565] The digital chassis 2510 includes a G-link interface circuit thatprovides outputs to a plurality of receiver LPAs 2520 via a plurality of32-bit busses. The LPAs 2520 perform the FFTs and channel estimation inparallel (e.g., one of the LPAs performs signal processing on each ofthe even symbols, while the other performs equivalent signal processingsteps on the odd receive symbols).

[0566] The LPAs 2520 provide the processed signals to LPAs 2530, thatare substantially similar in construction to the LPAs 2520 and the LPAs2110 and 2120. The LPAs 2530 perform QR decomposition and output thedecomposed signals to LPA cards 2540. The LPA cards 2540 perform matrixoperations involved in the null-steering and code-nulling procedures.The retrodirective weights calculated within the LPA cards 2540 areprovided as inputs to LPA cards 2550 in the transmitter path for useduring data spreading, beam forming, and generating IFFTs.

[0567] An additional LPA card 2560 is provided as a digital signalprocessing engine for the transmitter/receiver calibration (i.e., T/Rcompensation). The T/R calibration LPA card 2560 communicates with aprobe antenna 2565 via a G-link interface, ananalog-to-digital/digital-to-analog converter, and a transmit/receivecalibration module 2570. The transmit/receive calibration module 2570includes a receiver, a transmitter, a transmitting amplifier, and atransmit/receive switch. As described briefly above, the purpose of theprobe antenna is to compensate for distortion due to the transmitter andreceiver paths through the base station 110. That is, thetransmit/receive modules 2512 introduce a certain amount of distortionand phase delay into the transmitted and received signal so that it isnecessary to compensate for these distortions to provide an accurateproduction of the transmit and receive signals. The probe antenna pathacts like a remote station so that when the base station 110 istransmitting from the antenna array 120, this information is received onthe probe 2565. Conversely, when the probe antenna 2565 is transmitting,the antenna array 120 of the base station 110 is receiving the knownsignal transmitted by the probe antenna 2565. By signal processingperformed within the transmit/receive calibration LPA card 2560, thedifferential amplitude and phase across the phase transmitter andreceiver paths can be determined. Thus, the base station 110 cancompensate for these distortions by means of the signals transmitted andreceived by the probe antenna 2565.

[0568] A global positioning system antenna 2580 receives GPS timing toprovide a reference clock for each of the local oscillators within thebase station 110. This ensures that accurate synchronization can beobtained throughout the entire wireless communication system 100.

[0569]FIGS. 6 and 7 show alternative embodiments of the directionalantenna arrays 120 that may be used in the system of the presentinvention. A first embodiment of the base station antenna implementationis designated generally as 120 a. The antenna 120 a is a circular patchslot array antenna including a protective RADOME 505 available fromRADIX Technologies, Inc. of Mountain View, Calif., a generallycylindrical housing 507, and a support pole 510. A plurality ofmulti-element vertical patch arrays 515 are depicted in cutaway in FIG.6. Each of the patch arrays 515 are capable of directionally emittingradio frequency signals so as to provide beam forming capabilitiesnecessary for the proper implementation of the present invention. In oneembodiment, the height of the cylindrical portion 507 is approximately18″, while the diameter of the RADOME 505 is approximately 5-16″.

[0570] In one advantageous embodiment, the antenna 120 a includes avertical stack of 4 microstrip patch antennas. Four of these stacks willrespectively be oriented to cover four 90° quadrants. Thus, a total of16 circumferential stacks of microstrip flared-notch antennas (whereeach vertical stack comprises eight notches) will be included on thebase antenna 120 a. For both the remote and base antennas, the preferredsensor element spacing is one-half wavelength.

[0571]FIG. 7 depicts a second implementation of the base station antennaof the present invention that is generally designated as 120 b. Theantenna 120 b includes a RADOME 520, a generally cylindrical portion525, and a support pole 530. The RADOME 520 is approximately 18-24″ indiameter while the cylindrical portion 525 is approximately 14″ inheight. As shown in cutaway, the antenna 120 b includes a flaredcircular horn configuration 535 as well as a plurality of monopoletransmission elements 540. The monopole elements 540 may be used forbeam forming purposes such as that that is necessary for the optimumoperation of the present invention.

[0572]FIG. 80 is a transceiver block diagram showing the main structuralelements of the down converter 305 depicted in FIG. 72. As shown in FIG.80, the antenna 190 and the transmit/receive switch 300 connect tobandpass filters 702, 704 that, in turn, connect to amplifier 706, 708,respectively. The path through the filter 702 and the amplifier 706constitutes the receive path that is part of the down converter 305,while the path that is through the amplifier 708 and the bandpass filter704 constitutes part of the transmission pass that is a part of the upconverter circuit 307. The output of the amplifier 706 and the input ofthe amplifier 708 connect to a switch 710. The switch 710 is used toswitch between the transmission and receiving paths associated with thedown and up converters 305, 307, respectively.

[0573] Although the up converter 307 and the down converter 305 arerepresented in FIG. 72 as functionally distinct blocks, it will beappreciated by one of ordinary skill in the art that the same structuralelements may be used to perform the functions of both the up converterand the down converter in an architecture that reuses amplifiers and sawfilters within the transmitter and receiver path. The switch 710connects to a bandpass filter 712. In one advantageous embodiment, thebandpass filter 712 has a bandpass frequency between 1,865 MHz and 1,950MHz. The bandpass filter 712 connects to a multiplier 715 that receivesan input from a first local oscillator having an oscillation frequencyof 1667.5 MHz. The multiplier 715 connects to a digital attenuatorcircuit 720 that receives a gain control input from the demodulator 310(see FIG. 72). The digital attenuator 720 connects to an amplifier 724via a switching circuit 722. The switching circuit 722 allows theamplifier 724 to be used bidirectionally in both the transmitter andreceive paths. That is, when switched in a first direction, the outputof the amplifier 724 connects to the digital attenuator circuit 720while when switched in a second mode, the input of the amplifier 724connects to the digital attenuation circuit 720. By using the sameamplifier (i.e., the amplifier 724) in both the transmitter and receiverpaths the same amplifier characteristics are observed in both paths sothat transmission and reception compensation is greatly simplified. Theswitching network 722 further connects to a summing circuit 725.

[0574] When operating in a receiving mode, the summing circuit 725 actsas a signal splitter while, when in the transmitting mode, the summingcircuit 725 acts to linearly add a pair of input signals. The summingcircuit 725 connects to parallel amplification and filtering pathshaving corresponding elements. Specifically, one input to the summingcircuit 725 comprises a saw bandpass filter 730 having a centerfrequency of 270 MHz and a bandwidth of 1.5 MHz. A corresponding sawbandpass filter 732 has a center frequency of 200 MHz and a bandwidth of1.5 Mhz. The bandpass filters 730, 732 connect, respectively, toamplifiers 738, 740 via switching networks 734, 736. Again, theswitching networks 734, 736 insure that identical amplifiercharacteristics are observed in both the transmit and receive paths. Theamplifiers 738, 740 advantageously provide an amplification factor. Theswitching circuits 734, 736 connect to corresponding saw bandpassfilters 742, 744. The bandpass filter 742 has a center frequency ofapproximately 280 Mhz and a bandwidth of 1.5 Mhz, while the bandpassfilter 744 has a center frequency of 200 Mhz and a bandwidth of 1.5 Mhz.The bandpass filters 742, 744 connect, respectively, to correspondingamplifiers 750, 752 via switching networks 746, 748. The amplifiers 750,752 advantageously provide an amplification factor. The switchingnetworks 746, 748 connect to corresponding multipliers 754, 756. Themultiplier 754 receives a local oscillator input signal oscillating at281.25 Mhz, while the multiplier 756 receives a local oscillator inputsignal oscillating at approximately 201.25 Mhz.

[0575] The multipliers 754, 756 connect to corresponding low passfilters 758, 760, that in turn connect to switches 762, 764,respectively. The switch 762 receives an input signal from an amplifier766 and provides an output signal to an amplifier 768, while the switch764 receives an input signal from an amplifier 770 and provides anoutput signal to an amplifier 772. The amplifiers 766 through 772advantageously have an amplification factor. Amplifiers 766, 770 form apart of the transmission path, and therefore properly belong to the upconverter 307, while the amplifiers 768, 772 belong to the receptionpath and therefore, properly belong to the down converter 305 of FIG.72. The amplifiers 766, 770 connect to digital-to-analog converters 774,778, respectively. The digital-to-analog converters 774, 778 alsocomprise a portion of the up converter 307 and receive adigital-to-analog clock pulse from the synchronization circuit 312. Theamplifiers 768, 772 connect to analog-to-digital converters 776, 780,also comprise a portion of the down converter 305 that receiveanalog-to-digital converter clock inputs from the synchronizationcircuit 312 (see FIG. 72).

[0576] The inputs to the digital-to-analog converters 774, 778 arereceived from the modulation circuit 345, while the outputs of theanalog-to-digital converters 776, 780 are provided as inputs to thedemodulation circuit 310.

[0577] The operation of the up/down converter circuit depicted in FIG.80 will first be described with reference to the received path and willnext be described with reference to the transmission path. Within thereceived mode, signals picked up by the antenna 120 are transmitted tothe switch 300 and passed through the bandpass filter 702 so as toattenuate any signals that are not within the frequency band of interest(i.e., frequencies between 1,865 Mhz and 1,950 Mhz). The filteredsignals are then amplified by an amplification factor within theamplifier 706. The output of the amplifier 706 is provided as an inputto the switch 710 that allows the amplified signal to be passed throughthe bandpass filter 712 that further filters out any undesired signalsoutside of the designated bandpass range.

[0578] Signals that are allowed to pass through the filter 712 aremultiplied by the local oscillator frequency at 1,667.5 Mhz within themultiplier 715. Thus, the multiplier 715 acts as a synchronous detectorthat may be used to cause a first down conversion of the signal fromapproximately the 2 GHz range down to the 200 to 300 Mhz range. Thisdown-converted signal is then attenuated by means of the digitalattenuation circuit 720 and amplified with an amplification factor bymeans of the amplifier 724. The down-converted signal is then splitwithin the signal splitter 725 so that one portion of the signal entersthe saw bandpass filter 730 while an identical portion of the signalenters the saw bandpass filter 732.

[0579] The portion of the signal that enters the bandpass filter 730 isfiltered to attenuate signals outside of the frequency range of 279.25Mhz and 280.75 Mhz. This filtered signal is then amplified by a factorvia the amplifier 738 and is then filtered again through the filter 742having substantially identical characteristics to the filter 730. Onceagain, the filtered signal is amplified by the amplifier 750 with anamplification factor and this signal is input to the multiplier 754. Themultiplier 754 acts as a synchronous detector that converts the signaloutput by the amplifier 750 to substantially a base band signal bymultiplying the oscillator signal at 281.25 Mhz. The base band signal isthen passes through the low pass filter 758 and from there is suppliedas an input to the amplifier 768 via the switch 762. The amplifier 768amplifies the base band signal by a factor and this signal is thenconverted to digital data by means of the analog-to-digital converter776.

[0580] The second portion of the signal output by the splitter 725follows a substantially similar path to that followed by the firstportion of the signal output by the splitter 725, with the exceptionthat the second portion of the signal is filtered to pass bandwidthsbetween 199.25 Mhz and 200.75 Mhz. Furthermore, this portion of thesignal is synchronously detected within the multiplier 756 by means of alocal oscillator signal at 201.25 Mhz. In this manner, signals receivedby the antenna 120 are asynchronously detected, down-converted to thebase band level, and digitized so as to provide digital information tobe demodulated by the demodulator 310.

[0581] The transmission path for signals that are to be transmitted bythe high-bandwidth base station 110 is substantially the same throughthe up converter as through the down converter with the exception thatthe order of the signal processing steps is reversed. Specifically,modulated digital signals serve as the inputs to digital-to-analogconverters 774 and 778, so as to produce analog signals that areamplified by the amplifiers 766 and 770, respectively. The amplifiedanalog signals pass through the switching circuits 762, 764 and arefiltered by respective low pass filters 758, 760. Along the first paththe analog signal is up-converted by modulation (i.e., multiplication)with the local oscillator signal at 281.25 Mhz while the second signalis up-converted by modulation with an oscillator at 201.25 Mhz. Thefirst modulated signal is then amplified and filtered via the amplifiers750, 738 and the filters 742, 730 so as to provide a well defined signalbetween 200 and 79.25 Mhz and 280.75 Mhz. The second signal is likewiseamplified and filtered via the amplifiers 752, 740 and the filters 744,732, so as to provide a well defined signal within the frequency rangeof 199.25 Mhz and 200.75 Mhz. The two signals that are output from thebandpass filters 730 and 732 are provided as inputs to the summingcircuit 725. The summing circuit 725 linearly adds the two inputsignals, and these signals are amplified by the amplifier 724. Thedigital attenuation circuit 720 then attenuates the amplified outputsignal and the multiplier 715 further up converts this signal bymultiplication with the oscillator frequency at 1,667.5 Mhz. In thismanner, the original input signals containing the communicationinformation are up-converted to the transmission frequency range. Thesignal to be transmitted is then filtered between 1,865 and 1,950 Mhzwithin the filter 712 and the signals amplified in the amplifier 708after passing through the switch 710. The amplified transmission signalis further filtered within the bandpass filter 704 and this filtered andamplified signal is provided as an output to the antenna 120 via thetransmission/receive switch 300.

[0582]FIG. 80A is a schematic block diagram showing the main internalfunctional elements of the synchronization circuitry 312. As shown inFIG. 80A the synchronization circuit 312 includes a frequency controller785 that connects to a 40 Mhz reference oscillator 787 having a 2-bitinput from a data clock (not shown). The 40 Mhz reference oscillator 787outputs a signal to a divide-by-eight binary counter 789 that, in turn,supplies the output signal references for local oscillators 791, 793 and795. The local oscillator 791 provides the oscillation frequency at1,667.5 Mhz, while the oscillators 793, 795, respectively, provide theoscillation frequencies of 281.25 Mhz and 201.25 Mhz. Finally, thedivide-by-eight binary counter 789 further provides a clock input pulsefor each of the analog-to-digital and digital-to-analog converters 774through 780.

[0583]FIG. 81 depicts a schematic block diagram of the main elements ofthe down converter 405 within the base station 110 depicted in FIG. 73.Specifically, the antenna 120 connects to a bandpass filter 802 via thetransmit/receive switch 400 while the switch 400 is in the receive mode.The filter 802 passes frequencies about 1,865 Mhz and below 1,950 Mhz.The filter 802 connects to the input of an amplifier 804 that, in turn,connects to a second bandpass filter 806 that has substantially the samecharacteristics as a filter 802. The filter 806 provides an input to amultiplier 809 that also receives inputs from a local oscillator (notshown in FIG. 81) at an oscillation frequency of 1,667.5 Mhz. The outputof the multiplier 809 connects to a digital attenuator 811 that receivesa gain control input fee as the demodulator circuit 410 (see FIG. 73).The output ofthe digital attenuator 811 serves as the input to anamplifier 813 having an amplification factor.

[0584] The amplified signal output from the amplifier 813 enters asignal splitter 815 that divides the signal into, for example, sixsubstantially identical portions. Each of the six signals output by thesplitter 815 are filtered, amplified, down-converted and digitized insubstantially the same way.

[0585] The first signal enters a bandpass filter 817 having a centerfrequency of 281.5 Mhz with a bandwidth of 1.5 Mhz. The output of thebandpass filter 817 serves as an input to an amplifier 819 having anamplification factor. The output of the amplifier 819 serves as theinput to a bandpass filter 821 having substantially the samecharacteristics as a bandpass filter 817. The output of the bandpassfilter 821 connects to an amplifier 823 having an amplification factor,while the output of the amplifier 823 serves as the input to amultiplier 825. The multiplier 825 also receives a local oscillatorinput at 282.5 Mhz so as to act as a synchronous detection circuit thathas an output connected to a low pass filter 827. The output of the lowpass filter 827 serves as the input to an amplifier 829, while theoutput of the amplifier 829 serves as the input to an analog-to-digitalconverter 831. The analog-to-digital converter 831 further receives a 10Mhz clock input from the frequency reference circuit 409 (see FIG. 73).The output of the analog-to-digital converter 831 serves as the input tothe demodulator circuit 410 in FIG. 73.

[0586] The second portion of the signal output from the signal splitter815 is input to a saw bandpass filter 833 having a center pass frequencyof 280 Mhz and a bandwidth of 1.5 Mhz. The bandpass filter 833 connectsto the input of an amplifier 835 that, in turn, outputs a signal to abandpass filter 837 having substantially the same characteristics as thebandpass filter 833. The output of the filter 837 serves as the input toan amplifier 839 having an amplification factor. The output of theamplifier 839 connects as an input to a multiplier circuit 841 whichalso receives a local oscillator signal at 282.5 Mhz. The output of themultiplier circuit 841 serves as the input to a low pass filter 843that, in turn, connects to the input of an amplifier 845 having anamplification factor. The output of the amplifier 845 is input to ananalog-to-digital converter 847 that operates off of ananalog-to-digital clock of 10 Mhz. The 10 Mhz clock is received from thefrequency reference circuit 409 of FIG. 73. The output of theanalog-to-digital converter 847 serves as an input to the demodulatingcircuit 410 (see FIG. 73).

[0587] The third portion of the signal output by the signal splitter 815enters a bandpass filter 849 that has a center pass frequency of 278.5Mhz and a bandwidth of 1.5 Mhz. The output of the bandpass filter 849enters the input of an amplifier 851 having an amplification factor,while the output of the amplifier 851 connects to the input of abandpass filter 853 having substantially the same bandpasscharacteristics as the filter 849. The output of the filter 853 connectsto the input of an amplifier 855 that, in turn, connects to a multiplier857 that connects to an analog-to-digital converter 863 via a low passfilter 859 and an amplifier 861. The amplifier 855, the multiplier 857,the low pass filter 859, the amplifier 861, and the analog-to-digitalconverter 863 are substantially identical to the corresponding elements823, 825, 827, 829 and 831, and function in substantially the samemanner.

[0588] The fourth portion of the signal output from the signal splitter815 enters a bandpass filter 865 having a center bandpass frequency of201.5 Mhz and a bandwidth of 1.5 Mhz. The output of the bandpass filter865 serves as the input to an amplifier 866 having an output connectedto a bandpass filter 867 that has substantially identical filteringcharacteristics as the bandpass filter 865. The output of the bandpassfilter 867 connects to the input of an amplifier 868 having anamplification factor. The output of the amplifier 868 connects to amultiplier 869 that also receives a local oscillator frequency of 202.5Mhz. Thus, the multiplier 869 acts as a synchronous detector thatoutputs a down-converted base band signal to a low pass filter 870. Thelow pass filter 870 provides an input to an amplifier 871 having anamplification factor and the output of the amplifier 871 serves as theinput to an analog-to-digital converter 872 that receives a 10 MJhzanalog-to-digital converter clock from the frequency reference circuit409. The output of the analog-to-digital converter 872 serves as aninput to the demodulating circuit 410 (see FIG. 73).

[0589] The fifth and sixth portions of the signals output by the signalsplitter 815 are provided as inputs to analog-to-digital converters 880,888, respectively, via bandpass filters 873, 881, amplifiers 874, 882,bandpass filters 875, 883, amplifiers 876, 884, multipliers 877, 885,low pass filters 878, 886, and amplifiers 879, 887, respectively. Eachof the circuit elements between the splitter 815 and theanalog-to-digital converters 880, 888 are substantially identical totheir corresponding elements between the signal splitter 815 and theanalog-to-digital converter 872, with the exception that the bandpassfilters 873 and 875 have a center frequency of 200 Mhz and the bandpassfilters 881, 883 have a center pass frequency of 198.5 Mhz.

[0590] The operation of the down converter portion of the base station110 is substantially similar to that of the down converter portion ofthe high-bandwidth base station 110. Specifically, signals received bythe antenna 120 and switched to the receiving path by the switch 400 arefiltered and amplified by means of the filters 802, 806 and theamplifier 804. Subsequently, the signal is down-converted to a lowerfrequency band by synchronous detection within the multiplier 809. Afterthe first down conversion step, the signal is digitally attenuated bymeans of the attenuator 811 and then amplified by means of the amplifier813. The signal is then split into a plurality of substantiallyidentical signals that each follow a different detection path. Each ofthe detection paths is substantially identical, with the exception thateach path down converts the detected signal into a different base bandfrequency range. Thus, for example, the first portion of the splitsignal is filtered about a center frequency of 281.5 Mhz by the bandpassfilters 817, 821, and is amplified by the amplifiers 819, 823. Thisfiltered signal then is synchronously detected by-the multiplier 825 andconverted to base band. This base band signal is subsequently filtered,amplified and digitized within the low pass filter 827, the amplifier829, and the analog-to-digital converter 831. This sequence of detectionis substantially the same for each of the six signal portions output bythe signal splitter 815, with the exception that the bandpass filtersoperate at different centering frequencies and the local oscillatorsignals that serve as inputs to the various multipliers are differentfor the bottom three signal portions than for the top three signalportions.

[0591]FIG. 81A is a simplified schematic block diagram showing the maininternal components of the frequency reference circuit 409. As shown inFIG. 81A, the frequency reference circuit 409 includes a frequencycontrol circuit 890, a 40 Mhz reference oscillator 891, and adivide-by-four circuit 892. The divide-by-four circuit 892 providesoutputs to local oscillators 893, 894 and 895, as well as to each of theanalog-to-digital converter circuits and the digital-to-analog convertercircuits (see FIG. 82). The local oscillator 893 provides the 1,667.5Mhz output signal, while the local oscillators 894 and 895,respectively, provide the 281.25 and the 201.25 Mhz oscillator signals.

[0592]FIG. 82 is a schematic block diagram that shows the main internalcomponents of the up converter 407 along the transmission path of thebase station 110 (see FIG. 73). The antenna 120 connects to a bandpassfilter 902 via the switch 400 when the switch 400 is in the transmissionmode. The bandpass filter 902 allows frequencies between 1,865 Mhz and1,950 Mhz to pass. The bandpass filter 902 connects to the output of apower amplifier 904 having an amplification factor. The input of thepower amplifier 904 connects to a bandpass filter 906 having frequencypass characteristics that are substantially the same as the bandpassfilter 902. The input of the bandpass filter 906 connects to the outputof a multiplier 908 that receives a first input from a local oscillatorhaving an oscillation frequency of 1,667.5 Mhz and a second input from adigital attenuator circuit 910. The digital attenuator circuit 910receives gain control inputs from the modulation circuit 445 (FIG. 73).The input of the digital attenuator circuit 910 connects to the outputof a power amplifier 912 that, in turn, receives inputs from a summingcircuit 914. The summing circuit 914 receives, in one embodiment, sixseparate inputs that are linearly added within the summing circuit 914to provide an output to the amplifier 912. Each of the six inputs to thesumming circuit 914 connects to a bandpass filter having a 1.5 Mhzbandwidth. Specifically, the bandpass filters 920, 930, 940, 950, 960and 970 serve as inputs to the summing circuit 914. The bandpass filters920, 930, 940, 950, 960 and 970, respectively, have center passfrequencies of 281.5 Mhz 280 Mhz, 278.5 Mhz, 201.5 Mhz, 200 Mhz and198.5 Mhz. Each of the bandpass filters 920-970, respectively, connectto the outputs of amplifiers 921-971. The amplifiers 921-971,respectively, receive inputs from bandpass filters 922-972. The bandpassfilters 922-972 have substantially the same frequency pathcharacteristics as the bandpass filters 920-970. The bandpass filters922-972 each connect to outputs of amplifier circuits 923-973.

[0593] The amplifier circuits 923-973 connect to the outputs ofrespective multipliers 924-974. The multipliers 924, 934, 944 receivelocal oscillator input signals at an oscillation frequency of 282.5 Mhz,while the multipliers 954, 964, 974 receive local oscillator inputs at202.5 Mhz. Each of the multipliers 924-974 connect to corresponding lowpass filters 925-975. The low pass filters, in turn, receive inputs fromthe output of respective amplifiers 926-976. Finally, each of theamplifiers 926-976, respectively, receive inputs from digital-to-analogconverters 927-977. Each of the digital-to-analog converters 927-977receive digital-to-analog converter clock input signals at 10 Mhz fromthe output of the divide-by-four binary counter 892 (see FIG. 81A) andalso receive inputs from the modulator circuit 445 shown in FIG. 73.

[0594] In operation, modulated data signals serve as inputs to thedigital-to-analog converters 927-977. The digital-to-analog converters927-977 convert the modulated digital data signals into analog signalsthat are subsequently amplified by the amplifiers 926-976, and filteredby the low pass filters 925-975. The outputs of the low pass filters925-975 enter as one input of the multipliers 924-974, respectively. Thesecond inputs of the multipliers 924-974 receive local oscillator inputsat either 282.5 Mhz or 202.5 Mhz. Thus, the signals output from the lowpass filters 925-975 are up-converted to a first high frequency level.The up-converted signals output by the multipliers 924-974 aresubsequently amplified and filtered by means of the amplifiers 923-973and 921-971, and the filters 922-972 and 920-970. The outputs to thefilters 920-970 serve as inputs to the summing circuit 914, thatlinearly sums each of the signals applied at the six input terminals.

[0595] The summed output of the summing circuit 914 serves as an inputto the power amplifier 912. The output of the power amplifier 912 entersthe digital attenuator circuit 910 so as to fine tune the gain controlapplied to the signal output from the signal amplifier 912, and theoutput of the digital attenuator 910 serves as the first input to themultiplier 908. The second input of the multiplier 908 is the localoscillator signal at 1,667.5 Mhz. Thus, the multiplier 908 serves to upconvert the signal output from the digital attenuator 910 to thetransmission frequency of the base station 110. The output of themultiplier 908 is subsequently filtered and amplified by means of thefilters 902, 906 and the amplifier 904. Finally, the output of thefilter 902 serves as the input of the switch 400 in the transmit mode,that relays this up-converted and amplified signal to the antenna 120.

[0596] Method of Dynamically Allocating Bandwidth

[0597] The bandwidth allocation method performed by the bandwidth demandcontroller (see FIG. 74) is depicted in FIG. 83. The method begins in astart block 3300. Once the bandwidth allocation method begins,initialization functions are performed including, as shown in FIG. 83,determining if the number of antenna sensor elements has been changedsince the last use of the base station 110 or remote terminal 187, 192.For example, it may be desirable to provide a base station 110 or remoteaccess terminal 187, 192 with increased spatial resolution capability sothat the base or remote station can more accurately discriminate betweenincoming signals. In such a case, the base station 110 or remoteterminal 187, 192 would be deactivated while a new antenna is installedhaving a greater number of sensor elements that, as well known in theart, would provide a greater degree of directional discrimination orspatial division for that base station or remote. Once the installationof a new antenna is complete, then the installer reactivates the base110 or remote 187, 192 and, as indicated within a decision block 3305 atest is performed to determine if a number of antenna elements ischanged. If a number of antenna elements has changed, control passes toan activity block 3310 wherein the number of tones within a tone set isredefined (e.g., to a smaller number if the number of antenna elementsincreases) so that the matrices used to calculate the complex weightsapplied to the sensors and tones within a tone set maintain the samedimensionality. Thus, as discussed above, essentially the same SINR ispreserved while processing costs are not increased. After theinitialization such as performed within the activity block 3310, controlpasses to a decision block 3315 wherein a determination is made if a newuser is requesting bandwidth. If it was determined, however, within thedecision block 3305 that the number of antenna elements has not beenchanged, then the method passes immediately to the decision block 3315to the decision block 3305.

[0598] If it is determined within the decision block 3315, that a newuser has not requested bandwidth through the access channel, thencontrol passes to a subroutine block 3320 wherein the bandwidthassignments already allocated within the communications link aremodified, if necessary, to maximize the SINR. Control returns from thesubroutine block 3320 to the decision block 3315 until it is determinedthat a new user is requesting bandwidth over the control access channel.

[0599] When a new user requests bandwidth, control passes to an activityblock 3325 to determine how much bandwidth is requested. As discussedabove, the requested bandwidth is predicated upon the type of data thatis transmitted (e.g., voice, video, data, etc.) as well as thetransmitting device. For example, if an individual telephone unit istransmitted, then as few as 8 kilobits per second of bandwidth may berequested, while if a P-1 link connected to a PBX is requestingbandwidth. as much as 1 544 Mhz will be requested. In one embodiment,the requesting device transmits an initialization or identificationsignal that indicates to the remote station 187, 192 the bandwidthrequirements of the requested device.

[0600] Once the quantity of bandwidth requested is determined within theactivity block 3325, control passes to a decision block 3330 or adetermination is made if the communications channel has sufficient freebandwidth to accommodate the requesting unit. If it is determined thatthe channel does not have sufficient free bandwidth to accommodate theoptimum bandwidth requested by the new user, then control of the methodpasses to an arbitration phase wherein a determination is first madewithin a decision block 3335 if the user can use less bandwidth. If auser cannot operate with less bandwidth than requested, then the user isdisconnected and access is denied to the communications channel asindicated within activity block 3340. However, if it is determined thatthe user can operate with less bandwidth, then the base station 110 asksthe user, via the remote station 187, 192, for a lower bandwidthrequirement as indicated within an activity block 3345. Control thenreturns to the activity block 3325 wherein the quantity of requestedbandwidth is again determined. Of course, it will be understood, thatthe base station may present a suggested bandwidth that is allowable tothe user via the remote station 187, 192 if the user is sophisticatedenough to determine if such a suggested bandwidth would be sufficient toprovide normal operation of the user communication device.

[0601] However, if it is determined that the communication channel hassufficient free bandwidth to accommodate the requesting user, thencontrol passes from the decision block 3330 to a decision block 3350wherein a test is performed to determine if there are any free tonesets. That is, if there are any tone sets that have not yet beenallocated to other users within the region of the requesting remoteterminal 187, 192. If there are free tone sets within the region of therequesting room or terminal 187, 192, then control passes to an activityblock 3355 wherein one or more of the free tone sets is allocated to theuser for use in transmitting data from the remote associated with theuser to the base within the remote spatial cell. Control then passesfrom the activity block 3355 to an activity block 3360. If there isdetermined, however, within the decision block 3350 that there are nofree tone sets, then control passes instead to an activity block 3365wherein one or more currently used tone sets are allocated to the userfor the transmission of data between the remote 187, 192 and basestation 110. It is possible that multiple tone sets will be allocated toa requesting user if a very high bandwidth is requested by the user. Itshould also be noted here, that because the tone sets are grouped intofour approximate 1 Mhz bands that when multiple tone sets are allocatedto a single user to establish a separate communications channel, thesetone sets are typically within the same 1 Mhz band.

[0602] Once control passes from either the activity block 3355 or theactivity block 3365 to the activity block 3360, one or more codes (i.e.,spreading codes used to modulate the various tones within the allocatedtone set or tone sets) are allocated to the user making sure that thesame code (i.e., on the same tone set) is not used by a proximate remoteto the remote connected to the new user. In this manner, maximumfrequency and code reuse is achieved by spatially separating usershaving the same tone sets and code assignments. Of course, it will beunderstood, that due to the adapted channel equalization methoddescribed above, that the spreading codes initially assigned to theremote terminals that, on line, are typically not are well-definedcodes, but rather constitute linear adapted spreading weights tomaximize the SINR. Therefore, it is highly unlikely that a newlyallocated code will be identical to any of the spreading weightsassigned to remote terminals within the same proximity as the remoteterminal assigned to the new user. As discussed in greater detail above,the criteria for modifying spreading codes assigned to each new userrequires that the spreading weights be linearly independent to provideat least one degree of freedom for each user within a given spatial cellsite.

[0603] Control passes from the activity block 3360 to a decision block3370 wherein a determination is made if the maximum constellation size(i.e., for any arbitrary M-ary modulation format) is sufficient tomaintain the requested bandwidth given the number of tone sets and codesallocated to the user. That is, if the newly defined communicationchannel tolerates a sufficiently high constellation size then therequired bandwidth will be satisfied for the requesting user. However,if the channel is not always resistant enough to handle the necessaryconstellation size to maintain the bandwidth required for operation ofthe new user, then additional codes or tone sets must be allocated tothe user in accordance with the method described. Once the tone sets,codes, and modulation format are defined for the newly requestedcommunication channel, control passes to the subroutine block 3320wherein the bandwidth assignments are modified, as necessary, tomaximize the SINR. Control then returns to the decision block 3315 andthe process repeats as described.

[0604] Alternative Embodiment of the Invention: Adaptive Beamforming forPlural Discrete Tones Followed by Combining Resultant Signals

[0605]FIG. 84A and FIG. 84B show an alternate embodiment of theinvention, where the spectral processing and the spatial processing areseparated. The spatial weights are computed independently for eachcarrier frequency. The spatial weights are then multiplied by spectralweights which are again calculated separately to produce a compositeweight. In other words, the combined spectral/spatial beamformer isbroken into a separate spectral beamformer and a separate spatialbeamformer which operate independently. FIG. 84A shows how the receivedsignals on the antennas through M−1 are processed by the spatialbeamformer to produce the coefficients A0 through AN−1. FIG. 84A alsoshows how the received signals on the antennas 0 through M−1 areprocessed by the spectral beamformer to produce the coefficients B0through BN−1. FIG. 84B shows how the spatial coefficients A0 are appliedto tone frequency 0 and how the result thereof is then independentlyoperated on by the spectral coefficients B0, with the resultant signalsthen being transmitted on the antennas through M−1.

[0606] Similarly, FIG. 84B shows how the spatial coefficients AN−1 areapplied to tone frequency n−1 and how the result thereof is thenindependently operated on by the spectral coefficients BN−1, with theresultant signals then being transmitted on the antennas 0 through M−1.Thus, it is seen how the spatial weights are computed independently foreach carrier frequency and then the spatial weights are multiplied byspectral weights which are calculated separately to produce a compositeweight.

[0607] In one form of this alternate embodiment, the base station andthe remote unit can exchange as few as two tones, one in each of the twosub-bands. The separation of 80 Mhz between the two sub-bands spreadsthe tones far enough apart so that noise bursts and interfering signalsin one sub-band do not degrade the other in the other sub-band. The twotones can be separately processed by spatial spreading and despreadingand thereafter combined to form the resultant signal. This alternateembodiment has the advantage of a simplified computation, whileretaining a reasonable immunity to noise and interference.

[0608] At the receiving station, each tone received by the multielementantenna array is spatially despread in a process analogous to receivebeamforming. The resultant signals are then combined. A first method ofsignal-combining is equal gain combining, where the signals are addedtogether. An alternate method of signal combining is maximal rationcombining, where the output signal is chosen from the two tones havingthe better SINR.

[0609] At the transmitting station, the alternate embodiment spatiallyspreads a data signal modulated with the first tone. The spatialspreading uses spatial spreading codes in a process analogous totransmit beamforming. Separately, the alternate embodiment spatiallyspreads the data signal modulated with the second tone. Then, the twospatially spread signals are combined and transmitted from themulti-element antenna array, forming a transmitted spread signal that isspectrally and spatially spread.

[0610] The alternate embodiment of the invention can have the spatialdespreading steps adaptively position spatial directions of the receiversensitivity towards a desired signal source and/or diminish the receiversensitivity from interfering sources. The alternate embodiment can alsohave et spreading steps adaptively position transmitted signal energy ofthe transmitted despread signal towards a source of the received spreadsignal and/or adaptively diminish the transmitted signal energy towardsinterferers. The alternate embodiment works well within the TDDprotocol.

[0611]FIG. 85A is a flow diagram of the computational steps performed inthe base station. In the transmission portion of the base station,traffic symbols are input on line 5 to the smear matrix step 10. Linkmaintenance pilot signals are input on line 7 to the digital signalprocessor (DSP) data processing RAM 12. Stored pilot signals are outputfrom the RAM 12 to the link maintenance pilot (*LMP) register 14 and arethen applied as one input to the smear step 10. The smear matrix 16 isalso applied to the smear step 10. The output of the smear matrix 16 isalso applied to the smear step 10. The output of the smear step 10 isapplied to the gain emphasis step 20. The values from the gain RAM 25are applied to the gain emphasis step 20 to provide output values whichare then applied to the beam form spreading step 30. Spreading weightsin a spread weight RAM 32 are applied to the beam form spread step. TheX vector is output on line 40 from the beam form spread step and is sentto the transmitter for transmission to the remote stations.

[0612] On the receive side of the signal processing in the base station,the X vector from the receivers is input on line 50 to the beam formdespread step 60. The despreading weight RAM 62 applies the despreadingweights to the beam form despread step 60. The signal output from thebeam form despread 60 is then applied to the gain emphasis step 70.Values from the gain RAM 25 are applied to the gain emphasis step 70.Values form the gain RAM 25 are applied to the gain emphasis step 70.Values output from the gain emphasis step 70 are applied to the desmearstep 80. Values for pilot signals from the gain emphasis step 70 arestored in the LMP register 72 and are applied to the desmear step 80.The desmear matrix is also applied form step 74 to the desmear step 80.Traffic symbols output from the desmear step 80 on line 82 are thenavailable to be utilized and further distributed in the base station.The pilot signals output form the LMP register 72 are stored in the LMPdigital signal processing DP RAM 76 and are then output on line 78.

[0613] Various values used in the spreading and despreading computationsare updated as is shown in FIG. 85A. The X vector input on line 50 isapplied to the updated weight step 54. The X vector input online 50 isalso applied to the data correction step 93 whose output is applied tothe update weight step 54. The updated weight values output from theupdated weight step 54 are sent to the valid weights step 56 and arethen output to the despread RAM 62. The traffic establishment support 86provides values to the property map 84 which processes traffic signalsfrom line 82 and applies the output to the smear step 89. Maintenancepilot signals online 81 are applied to the digital signal processing DPRAM 83 whose output is applied to the LMP register 85 whose output isapplied to the smear step 89. The smear matrix 87 is also applied to thesmear step 89. The output of the smear step 89 is applied to the gaindeemphasis step 91 whose output is applied to the data correlation step93 whose output is applied to the updated weight step 54 as previouslydescribed. In addition, the output from the smear step 89 is applied tothe element-wise gain covariance step 64. Another input to theelement-wise gain covariance step 64 is applied from the output of thebeam form despread step 60. The output of the element-wise gaincovariance step 64 is applied to the block normalization of elementsstep 66 which is in turn applied to the element-wise conjugation step 68which outputs the values to the gain RAM 25. In this manner, the basestation can perform both despreading operations for received signalvectors online 50 and spreading operations to transmit traffic symbolsinput on line 5, in accordance with the invention.

[0614]FIG. 85B shows the processing of the common access channelsignals. Two common access channels (CAC) signals from the transmitterare processed. A first signal is processed being received on the inputline 102 and is applied to the RMGS auto-correlation step 104, whoseoutput goes to the digital signal matrix step 106 whose output goes tothe digital signal processor. The common access channel signal online102 is also applied to the select ungated packets step 108 and to theselect gated packets step 110. The output of the select ungated packets108 is applied to the subtract even/odd packets step 112. The output ofthe selected gate packets 110 is applied to the apply code key step 114.The CAC code key step 116 applies it's value to the apply code key step114. The output of the apply code key step 114 is also applied to thesubtract even/odd packets step 112. The output of the subtract even/oddpackets step 112 ids applied to the RMGS autocorrelation step 118, whoseoutput is also applied to the compute T matrix step 106. The output ofthe compute T matrix step 106 is then applied to the digital signalprocessor.

[0615] A second one of the two CAC signals input from the receiver online 120 is applied to the select ungated packets step 122 and theselect gated packets step 124. The output of the select ungated packetsstep 122 is applied as one input to the combined gated/ungated packetsstep 126. The output of the selected gated packet steep 124 is appliedto the apply code key step 128 which also receives a signal from the CACcode key step 130. The output of the applied code key step 128 is thesecond input to the combined gated/ungated packets step 126. The outputof the combined gated/ungated packets step 126 is applied to the applydespread weight step 132. A signal from the digital signal processor isapplied to the rotated weight RAM 134 whose output is applied to thecompute despread weight step 136. The output of the compute despreadweight step 136 is applied to the apply despread weight 132, whoseoutput is sent to the digital signal processor. In this manner, thesteps shown in FIG. 85B carry out processing of the common accesschannel signals.

[0616] Although the preferred embodiments of the invention have beendescribed in detail above, it will be apparent to those of ordinaryskill in the art that obvious modifications may be made to the inventionwithout departing from its spirit or essence. For example, signalconstellation formats other than PSK, BPSK and QAM could be used inaccordance with the system of the present invention. Furthermore, thesystem could optimize for bit error rate (BER) rather than SINR. Also,the number of tones in a tone set, and the number of tone sets andcluster sets in a band could be selected based upon the specificapplication. The selected frequency bands could also be varied as calledfor by specific conditions. The TDD format could be altered based uponthe multipath environment to insure that an effectively static channelis observed in successive TDD frames. The maximization of the SINR couldbe performed based upon some signal property other than constantmodulus, etc. Consequently, the preceding description should be taken asillustrative and not restrictive, and the scope of the invention shouldbe determined in view of the following claims.

What is claimed is:
 1. A highly bandwidth-efficient communicationsmethod, comprising: receiving a received spread signal at a base stationhaving a multi-element antenna array with a first plurality of antennaelements arranged in a spaced vertical direction and a second pluralityof antenna elements arranged in a spaced horizontal direction; saidreceived spread signal comprising a first data signal spread over aplurality of discrete tones in accordance with a remote spreading codeassigned to a remote unit for a first time period; adaptivelydespreading the signal received at the base station by using firstdespreading codes that are based on the characteristics of the receivedsignals at the first plurality of antenna elements of said array andperform vertical beam steering; and adaptively despreading the signalreceived at the base station by using second despreading codes that arebased on the characteristics of the received signals at the secondplurality of antenna elements of said array to perform horizontal beamsteering.
 2. The highly bandwidth-efficient communications method ofclaim 1, further comprising: spreading a second data signal at the basestation with first spreading codes derived from said first despreadingcodes, that distributes the second data signal over a plurality ofdiscrete tones and the first plurality antenna elements of said array,forming a first spectrally spread signal that is spectrally andspatially spread vertically; spreading the second data signal at thebase station with second spreading codes derived from said seconddespreading codes, that distributes the second data signal over theplurality of discrete tones and the second plurality antenna elements ofsaid array, forming a second spectrally spread signal that is spectrallyand spatially spread horizontally; and transmitting said first andsecond spread signals during a second time period.
 3. The highlybandwidth-efficient communications method of claim 1, furthercomprising: receiving at the base station during an initializationperiod, a pilot spread signal comprising a known data signal spread overa plurality of discrete tones; correlating the known data signal fromthe pilot spread signal with a reference known data signal and formingsaid first despreading code that is based on the characteristics of thereceived signals at the first plurality of antenna elements arranged inthe spaced vertical direction, where a given element of the firstdespreading code corresponds to a given one of the first antennaelements and a given one of the discrete tones; and correlating a knowndata signal from the pilot spread signal with a reference known datasignal and forming said second despreading code that is based on thecharacteristics of the received signals at the second plurality ofantenna elements arranged in the spaced horizontal direction, where agiven element of the second despreading code corresponds to a given oneof the second antenna elements and a given one of the discrete tones. 4.The highly bandwidth-efficient communications method of claim 2, whereinboth the first and the second spread signals have a spectral form of adiscrete multitone signal transmitted on multiple elements in the array.5. The highly bandwidth-efficient communications method of claim 1,wherein the despreading step is a multiplication of a complex numberrepresentation of the despreading codes times a complex numberrepresentation of the received spread signal.
 6. The highlybandwidth-efficient communications method of claim 2, wherein thespreading step is a multiplication of a complex number representation ofthe second spreading codes times a complex number representation of thedata signal to be transmitted.
 7. The highly bandwidth-efficientcommunications method of claim 1, wherein the despreading stepsdetermine values of complex despreading codes which are then multipliedwith a complex number representation of the received signals, resultingin an estimate of the first data signal.
 8. The highlybandwidth-efficient communications method of claim 1, wherein thedespreading steps adaptively position the spatial direction of receivesensitivity towards a desired signal source and diminish receivesensitivity from interfering sources
 9. The highly bandwidth-efficientcommunications method of claim 2, wherein the spreading steps adaptivelyposition transmitted signal energy of the spread signals towards asource of the received spread signal and adaptively diminishestransmitted signal energy towards interferers.
 10. The highlybandwidth-efficient communications method of claim 2, wherein the firstperiod and the second period are parts of a time division duplex period.11. The highly bandwidth-efficient communications method of claim 1,wherein the antenna array has a planar symmetry.
 12. The highlybandwidth-efficient communications method of claim 1, wherein theantenna array has a cylindrical symmetry.
 13. A highlybandwidth-efficient communications system, comprising: means forreceiving a received spread signal at a base station having amulti-element antenna array with a first plurality of antenna elementsarranged in a spaced vertical direction and a second plurality ofantenna elements arranged in a spaced horizontal direction; saidreceived spread signal comprising a first data signal spread over aplurality of discrete tones in accordance with a remote spreading codeassigned to a remote unit for a first time period; means for adaptivelydespreading the signal received at the base station by using firstdespreading codes that are based on the characteristics of the receivedsignals at the first plurality of antenna elements of said array andperform vertical beam steering; and said despreading means adaptivelydespreading the signal received at the base station by using seconddespreading codes that are based on the characteristics of the receivedsignals at the second plurality of antenna elements of said array toperform horizontal beam steering.
 14. The highly bandwidth-efficientcommunications system of claim 13, further comprising: means forspreading a second data signal at the base station with first spreadingcodes derived from said first despreading codes, that distributes thesecond data signal over a plurality of discrete tones and the firstplurality antenna elements of said array, forming a first spectrallyspread signal that is spectrally and spatially spread vertically; saidmeans for spreading also spreading the second data signal at the basestation with second spreading codes derived from said second despreadingcodes, that distributes the second data signal over the plurality ofdiscrete tones and the second plurality antenna elements of said array,forming a second spectrally spread signal that is spectrally andspatially spread horizontally; and means for transmitting said first andsecond spread signals during a second time period.
 15. The highlybandwidth-efficient communications system of claim 13, furthercomprising: means for receiving at the base station during aninitialization period, a pilot spread signal comprising a known datasignal spread over a plurality of discrete tones; means for correlatingthe known data signal from the pilot spread signal with a referenceknown data signal and forming said first despreading code that is basedon the characteristics of the received signals at the first plurality ofantenna elements arranged in the spaced vertical direction, where agiven element of the first despreading code corresponds to a given oneof the first antenna elements and a given one of the discrete tones; andsaid correlating means also correlating a known data signal from thepilot spread signal with a reference known data signal and forming saidsecond despreading code that is based on the characteristics of thereceived signals at the second plurality of antenna elements arranged inthe spaced horizontal direction where a given element of the seconddespreading code corresponds to a given one of the second antennaelements and a given one of the discrete tones.
 16. The highlybandwidth-efficient communications system of claim 14, wherein both thefirst and the second spread signals have a spectral form of a discretemultitone signal transmitted on multiple elements in the array.
 17. Thehighly bandwidth-efficient communications system of claim 13, whereinthe despreading is a multiplication of a complex number representationof the despreading codes times a complex number representation of thereceived spread signal.
 18. The highly bandwidth-efficientcommunications system of claim 14, wherein the spreading is amultiplication of a complex number representation of the secondspreading codes times a complex number representation of the data signalto be transmitted.
 19. The highly bandwidth-efficient communicationssystem of claim 13, wherein the despreading determines values of complexdespreading codes which are then multiplied with a complex numberrepresentation of the received signals, resulting in an estimate of thefirst data signal.
 20. The highly bandwidth-efficient communicationssystem of claim 13, wherein the despreading adaptively positions thespatial direction of receive sensitivity towards a desired signal sourceand diminishes receive sensitivity from interfering sources.
 21. Thehighly bandwidth-efficient communications system of claim 14, whereinthe spreading adaptively positions transmitted signal energy of thespread signals towards a source of the received spread signal andadaptively diminishes transmitted signal energy towards interferers. 22.The highly bandwidth-efficient communications system of claim 14,wherein the first period and the second period are parts of a timedivision duplex period.
 23. The highly bandwidth-efficientcommunications system of claim 13, wherein the antenna array has aplanar symmetry.
 24. The highly bandwidth-efficient communicationssystem of claim 13, wherein the antenna array has a cylindricalsymmetry.